Pulse delay circuits. Square pulse delay device

We will begin our consideration of methods for spectral analysis of radio signals with deterministic periodic signals. As already emphasized above, deterministic signals are characterized by the fact that in any advance at the moment time, its value can be accurately determined. Periodic deterministic signal is a signal of a known shape that repeats periodically after an interval of time called a repetition period. Mathematically, a periodic signal is described by the expression

, (2.1)

Periodic signals include a harmonic oscillation defined over an infinite time interval, a sequence of pulses with a known amplitude, duration and repetition period, and others.

Spectral analysis involves choosing a system of basis functions. In practice, trigonometric functions are most widely used. This is due to the fact that when converting signals of this form, for example, linear radio circuits their shape is preserved, and only the amplitude and phase of the oscillations change. On the other hand, the formation of such signals is carried out by fairly simple technical means.

Signals described by trigonometric functions are called harmonic signals, and spectral analysis in the system of basic trigonometric functions – harmonic analysis.

So, we choose as basis functions the system

It is easy to verify that the functions forming system (2.2) are orthogonal on the time interval and satisfy the periodicity condition (2.1). Then, in accordance with (1.36), we can write

Where .

The norms of the basis functions in accordance with (1.26) are equal to

; .

Then from (1.39) it follows

, (2.4)

, . (2.5)

Expression (2.3) is called trigonometric Fourier series and represents the decomposition of the signal into components in a system of trigonometric functions.

In radio engineering practice, a different representation of series (2.3) often turns out to be more convenient. Let us select the kth component from (2.3)

and present it in the form

, (2.6)

From a geometric point of view, a component can be considered as a vector in the coordinate system (Fig. 2.1). The length of the vector, and is the angle by which the vector is rotated relative to the axis. It is easy to verify that

, .

Then expression (2.6) takes the form

Where .

Taking into account (2.7), the Fourier series (2.3) can be rewritten as follows

. (2.8)

Component

(2.9)

called k-th harmonic component or just k-th harmonic.

In accordance with the definition of spectrum given in the previous section, the collection and make up amplitude spectrum, and the totality – phase spectrum signal. Thus, the amplitude spectrum periodic signal contains a constant component and an infinite number of amplitudes of the corresponding harmonics. The same applies to the phase spectrum.

In spectral analysis, it is convenient to represent spectra in the form spectral diagrams.

Figure 2.2, a shows a periodic signal in coordinates and . Let's draw another axis perpendicular to the and axes and plot the values ​​on this axis. Let us depict the harmonic components of the signal at these frequencies, and plot the values ​​on the frequency axis in the form of straight line segments. If we now rotate the entire coordinate system around the axis by 90º in the direction of the arrow, we will obtain a diagram of the amplitude spectrum of the signal (Fig. 2.2, b). In the same way, you can construct a spectral diagram of the phase spectrum, approximate view which is shown in Fig. 2.2, c.

2.2. Amplitude and phase spectra of a periodic sequence of rectangular pulses

As an example, we give the Fourier series expansion of a periodic sequence of rectangular pulses with amplitude, duration and repetition period, symmetrical about zero, i.e.

, (2.10)

Here

Expanding such a signal into a Fourier series gives

, (2.11)

where is the duty cycle.

To simplify the notation, you can enter the notation

, (2.12)

Then (2.11) will be written as follows

, (2.13)

In Fig. 2.3 shows a sequence of rectangular pulses. The spectrum of the sequence, as well as any other periodic signal, is discrete (line) in nature.

The spectrum envelope (Fig. 2.3, b) is proportional. The distance along the frequency axis between two adjacent spectrum components is , and between two zero values ​​(the width of the spectrum lobe) is . The number of harmonic components within one lobe, including the zero value on the right in the figure, is , where the sign means rounding to the nearest integer, less (if the duty cycle is a fractional number), or (if the duty cycle is an integer value). As the period increases, the fundamental frequency decreases, the spectral components in the diagram move closer together, and the amplitudes of the harmonics also decrease. In this case, the shape of the envelope is preserved.

When solving practical problems spectral analysis Instead of angular frequencies, cyclic frequencies are used, measured in Hertz. Obviously, the distance between adjacent harmonics in the diagram will be , and the width of one spectrum lobe will be . These values ​​are presented in parentheses in the chart.

In practical radio engineering, in most cases, instead of the spectral representation (Fig. 2.3, b), spectral diagrams of the amplitude and phase spectra are used. The amplitude spectrum of a sequence of rectangular pulses is shown in Fig. 2.3, c.

Obviously, the envelope of the amplitude spectrum is proportional to .

As for the phase spectrum (Fig. 2.3, d), it is believed that the initial phases of the harmonic components change abruptly by the amount -π when the sign of the envelope changes sinc kπ/ q. The initial phases of the harmonics of the first lobe are assumed to be zero. Then the initial phases of the harmonics of the second lobe will be φ = -π , third petal φ = -2π etc.

Let's consider another Fourier series representation of the signal. To do this, we use Euler’s formula

.

In accordance with this formula k-th component (2.9) of the signal expansion into a Fourier series can be represented as follows

; . (2.15)

Here the quantities and are complex and represent the complex amplitudes of the spectrum components. Then the series

Fourier (2.8) taking into account (2.14) will take the following form

, (2.16)

, (2.17)

It is easy to verify that expansion (2.16) is carried out in terms of the basis functions , which are also orthogonal on the interval, i.e.

Expression (2.16) is complex form Fourier series, which extends to negative frequencies. The quantities and , where denotes the complex conjugate of the quantity, are called complex amplitudes spectrum Because is a complex quantity, it follows from (2.15) that

AND .

Then the totality constitutes the amplitude spectrum, and the totality constitutes the phase spectrum of the signal.

In Fig. Figure 2.4 shows a spectral diagram of the spectrum of the sequence of rectangular pulses discussed above, represented by a complex Fourier series

The spectrum also has a line character, but unlike the previously considered spectra, it is determined both in the region of positive and in the region of negative frequencies. Since is an even function of the argument, the spectral diagram is symmetrical about zero.

Based on (2.15), we can establish a correspondence between the coefficients and expansion (2.3). Because

AND ,

then as a result we get

. (2.18)

Expressions (2.5) and (2.18) allow you to find the values ​​in practical calculations.

Let us give a geometric interpretation of the complex form of the Fourier series. Let us select the kth component of the signal spectrum. In a comprehensive k-i form the component is described by the formula

, (2.19)

where and are determined by expressions (2.15).

In the complex plane, each of the terms in (2.19) is represented as vectors of length , rotated at an angle and relative to the real axis and rotating in opposite directions with frequency (Fig. 2.5).

Obviously, the sum of these vectors gives a vector located on the real axis whose length is . But this vector corresponds to the harmonic component

As for the projections of vectors onto the imaginary axis, these projections have equal length, but opposite directions add up to zero. This means that signals presented in complex form (2.16) are actually real signals. In other words, the complex form of the Fourier series is mathematical an abstraction that is very convenient for solving a number of problems of spectral analysis. Therefore, sometimes the spectrum defined by the trigonometric Fourier series is called physical spectrum, and the complex form of the Fourier series is mathematical spectrum.

And in conclusion, we will consider the issue of energy and power distribution in the spectrum of a periodic signal. To do this, we use Parseval’s equality (1.42). When the signal is expanded into a trigonometric Fourier series, expression (1.42) takes the form

.

Constant component energy

,

and the energy of the kth harmonic

.

Then the signal energy

. (2.20)

Because average power signal

then taking into account (2.18)

. (2.21)

When the signal is expanded into a complex Fourier series, expression (1.42) takes the form

Where - energy of the kth harmonic.

The signal energy in this case

,

and its average power

.

From the above expressions it follows that the energy or average power of the k-th spectral component of the mathematical spectrum is half as much as the energy or power of the corresponding spectral component of the physical spectrum. This is due to the fact that the physical spectrum is distributed equally between the mathematical spectrum.

Expressions (2.20) – (2.12) allow you to calculate and construct spectral diagrams of energy or power distribution, i.e. energy spectra of a periodic signal.

2.3. Integral Fourier transform

The harmonic analysis of periodic signals discussed above can be generalized to non-periodic (single) signals. Let's return to a periodic signal of arbitrary shape (Fig. 2.6, a).

Let's increase the value to . Signals adjacent to the central one will shift to the right and left along the time axis. If we now direct, only a single signal of finite duration will remain on the time diagram (Fig. 2.6, b). If the signal power is different from zero, then the energy of such a signal is finite. Mathematically, this condition is equivalent to the requirement for the convergence of the integral

,

where is the absolute value of the function.

In other words, the function must be absolutely integrable.

Let's turn to the spectral diagrams (Fig. 2.2, b, c). Because the distance along the frequency axis between adjacent components is equal to

, (2.24)

then with increase the value decreases and the spectral components come closer together. In this case, the values ​​of the complex amplitudes of the components decrease. When the magnitude and spectrum changes from line to solid and represents an infinitely large number of harmonics and infinitesimal amplitudes.

Let us use the complex form of the Fourier series (2.16). Substituting expression (2.17) into this formula, we obtain

.

Then, taking into account the fact that and , we write

. (2.25)

Because in the limit at the value , then, in accordance with (2.24), turns into an infinitesimal increment , and the frequency of the kth harmonic turns into the current frequency . In this case, the limits of the internal integral in (2.25) expand from to , and the summation turns into an integration operation. Taking this into account, expression (2.25) takes the following form:

. (2.26)

The integral enclosed in parentheses of expression (2.26) describes complex spectrum single signal

. (2.27)

Then, taking into account (2.27), expression (2.26) will be written as follows

. (2.28)

Expressions (2.27) and (2.28) are respectively direct and inverse Fourier transform.

Let us find out the physical meaning of the complex spectrum of a single signal. Let's fix a certain frequency. Since for a periodic signal , then to calculate the complex amplitude in expression (2.17), the limits of integration can be extended to the region, i.e.

. (2.29)

On the other hand, at the same frequency for a single signal in accordance with (2.27)

. (2.30)

Since the integrals in (2.29) and (2.30) coincide, we can write

, (2.31)

here the period according to (2.24) is equal to

where is the elementary frequency interval, measured in hertz.

.

In practical radio engineering, the amplitude spectrum is often used instead of the complex spectrum. In this case

. (2.32)

It follows that it characterizes distribution density amplitudes of the components of the continuous spectrum of a single signal by frequency. If is a time-varying voltage or current, then the dimension is or .

Let us write (2.32) taking into account (2.24) in the form

. (2.33)

It follows that the envelope of the continuous spectrum of a single signal and the envelope of the corresponding periodic signal coincide in shape and differ only in scale. In practice, in a number of cases, when calculating the spectrum of a periodic signal, it is much easier to first find a single signal, and then, using relation (2.33), move on to the spectrum of the periodic signal.

Fourier transforms (2.27) and (2.28) are presented in complex form. Using the known relations

, (2.34, a)

, (2.34,b)

you can get the trigonometric form of the transformations. Thus, taking into account (2.34, b), expression (2.27) takes the following form

where the first integral is the real part, and the second is the imaginary part, i.e.

, (2.36)

. (2.37)

Then the modulus or amplitude spectrum is calculated by the formula

and the argument or phase spectrum - in accordance with the expression

. (2.39)

If the signal is even function of time, then the second integral in (2.35) is equal to zero, because the product is an odd function, and the limits of integration are symmetrical about zero. In this case it is described real and even function

If the signal is odd function of time, then the first integral vanishes and represents an odd and purely imaginary frequency function, i.e.

. (2.41)

Thus, (2.35), (2.40) and (2.41) characterize the trigonometric form of the direct Fourier transform.

Let us now turn to the inverse Fourier transform (2.28).

Considering that

expression (2.28) can be represented in the following form

,

or, in accordance with (2.34,a)

If is an even function, then the second integral is an odd function and its value is equal to zero. Then we’ll finally write down

As an example, consider the Fourier transform of a rectangular pulse with duration and amplitude defined over the interval

Using expression (2.27), after simple transformations we obtain

.

In Fig. Figure 2.7 shows the shape of the pulse and its spectral function.

Comparison of spectral diagrams in Fig. 2.4 and fig. 2.7b shows that the shapes of the envelopes of the line and continuous spectra coincide, which confirms the conclusions made earlier. In this case, both the envelope of the line and the envelope of the continuous spectra reach zero at frequencies ω = 2lπ/τ , Where . When the value of the spectral function is equal to the pulse area.

Let's move on to consider the basic properties of the Fourier transform. For brevity, we will symbolically represent a pair of transformations (direct and inverse) as follows:

1. Linearity of the Fourier transform

where and are arbitrary numerical coefficients.

Proving formula (2.43) is not difficult; to do this, it is enough to substitute the sum into expression (2.27).

2. Time shift property (delay theorem)

Because , then (2.44) can be represented as

Thus, a time delay of a signal by an amount leads to a change in its phase spectrum by .

3. Changing the time scale

. (2.46)

Depending on the value, either compression or stretching of the signal occurs in time. From (2.46) it follows that when a signal is compressed in time by a factor, its spectrum expands by the same factor. And vice versa.

4. Differentiation operation

. 2.47)

When differentiating a signal, all harmonic components of its spectrum change the initial phase by .

5. Integration operation

. (2.48)

When integrating a signal, all harmonic components of its spectrum change the initial phase by . Property (2.48) is valid if

6. If , That

The integral on the right side of expression (2.49) is called convolution. Thus, the Fourier transform of a product of signals is a convolution (with a coefficient) of their spectra. In the special case when and two signals are equal you can get the following relation:

which is the integral form of the Parseval equality (2.22). From this relationship it follows that the total energy of a non-periodic signal is equal to the sum of the energies of all its spectral components. At the same time, dependence

, (2.51)

represents spectral density energy or energy spectrum single signal.

2.4. Effective duration and effective signal width

To solve practical problems in radio engineering, it is extremely important to know the values ​​of the duration and width of the signal spectrum, as well as the relationship between them. Knowing the duration of a signal allows us to solve problems of efficient use of the time available for transmitting messages, and knowledge of the spectrum width allows us to effectively use the radio frequency range.

Solving these problems requires a strict definition of the concepts “effective duration” and “effective spectrum width”. In practice, there are a large number of approaches to determining duration. In the case where the signal is limited in time (finish signal), as is the case, for example, for a rectangular pulse, determining the duration does not encounter difficulties. The situation is different when the signal theoretically has an infinite duration, for example, an exponential pulse

In this case, the time interval during which the signal value can be taken as the effective duration. In another method, the time interval during which . The same can be said with regard to determining the effective spectral width.

Although in the future, some of these methods will be used in the analysis of radio signals and circuits, it should be noted that the choice of method significantly depends on the shape of the signal and the structure of the spectrum. So, for an exponential pulse, the first of these methods is more preferable, and for a bell-shaped signal, the second method is more preferable.

A more universal approach is that using energy criteria. With this approach, the effective duration and effective spectrum width are considered, respectively, the time interval and frequency range within which the overwhelming majority of the signal energy is concentrated

, (2.52)

, (2.53)

where is a coefficient showing how much of the energy is concentrated in the intervals or . Usually the value is chosen within .

Let us apply criteria (2.52) and (2.53) to determine the duration and width of the spectrum of rectangular and exponential pulses. For a rectangular pulse, all the energy is concentrated in the time interval or, therefore its duration is . As for the effective spectrum width, it was found that more than 90% of the pulse energy is concentrated within the first lobe of the spectrum. If we consider the one-way (physical) spectrum of the pulse, then the width of the first lobe of the spectrum is in circular frequencies or in cyclic frequencies. It follows that the effective width of the spectrum of a rectangular pulse is equal to

Let's move on to the definition of exponential momentum. The total pulse energy is

.

Using (2.52), we obtain

.

By calculating the integral on the left side of the equation and solving it, we can come to the following result

.

We find the spectrum of the exponential pulse using the Fourier transform

,

whence follows

.

Substituting this expression into (2.53) and solving the equation, we obtain

.

Let us find the product of the effective duration and the effective spectrum width. For a rectangular pulse this product is

,

or for cyclic frequencies

.

For exponential momentum

Thus, the product of the effective duration and the effective width of the spectrum of a single signal is a constant value that depends only on the shape of the signal and the value of the coefficient. This means that as the signal duration decreases, its spectrum expands and vice versa. This fact has already been noted when considering property (2.46) of the Fourier transform. In practice, this means that it is impossible to form short signal, having a narrow spectrum, which is a manifestation of physical uncertainty principle.

2.5. Spectra of non-integrable signals

One of the conditions for the applicability of the Fourier transform of a function describing the shape of a signal is its absolute integrability, which means the finite energy of the signal. At the same time, in a number of cases, spectrally satisfying this condition. This can be a harmonic oscillation used as a carrier oscillation during the modulation operation, signals described by a unit function, etc. However, the Fourier transform apparatus can be extended to these signals.

Let us first consider a signal of the form

Obviously such a signal has infinite energy. Let us formally apply the Fourier transform (2.27) to this signal

.

,

then (2.54) can be rewritten as follows

.

Using the table integral

,

where is the function discussed above.

Then, taking this expression into account, we get

From (2.55) it follows that the spectrum of a harmonic vibration defined over the time interval is equal to zero at all frequencies except and . At these frequencies, the value of the spectral components goes to infinity (Fig. 2.8, a)

If we put , which corresponds to a constant signal, then from (2.55) it follows

.

Thus, the spectrum of a constant signal is different from zero only at (Fig. 2.8, b). At this frequency the value of the spectral component is equal to infinity.

It can be shown [L.3] that the spectrum of a step signal

,

.

From the above it follows that the spectra of non-integrable signals can be calculated using the Fourier transform using the mathematical abstraction - function. Then the question arises: what is the spectrum of a signal, the shape of which is described by a function, i.e.

.

Applying (2.27) to this signal and taking into account the filtering property of the -function, we obtain

Consequently, a signal that is a product of a - function (in practice, a very short pulse of very large amplitude) has a uniform spectrum over the entire frequency range. This conclusion, important for radio engineering problems, will be used in the future.

2.6. Correlation-spectral analysis of deterministic signals

In many radio engineering problems, there is often a need to compare a signal and its copy, shifted by some time. In particular, this situation occurs in radar, where the pulse reflected from the target arrives at the receiver input with a time delay. Comparison of these signals with each other, i.e. Establishing their relationship during processing allows one to determine the parameters of target movement.

For quantification the relationship between the signal and its time-shifted copy, a characteristic is introduced

, (2.57)

Which is called autocorrelation function(AKF).

To explain the physical meaning of the ACF, we give an example where the signal is a rectangular pulse with duration and amplitude . In Fig. 2.9 shows a pulse, its copy shifted by a time interval and the product . Obviously, integrating the product gives the value of the area of ​​the pulse, which is the product . This value, when fixed, can be represented by a point in coordinates. When changing, we will get a graph of the autocorrelation function.

Let's find an analytical expression. Because

then substituting this expression into (2.57), we get

. (2.58)

If you shift the signal to the left, then using similar calculations it is easy to show that

. (2.59)

Then combining (2.58) and (2.59), we get

. (2.60)

From the example considered, the following important conclusions can be drawn that apply to arbitrary waveforms:

1. The autocorrelation function of a non-periodic signal decreases with growth (not necessarily monotonically for other types of signals). Obviously, the ACF also tends to zero.

2. The ACF reaches its maximum value at . In this case, it is equal to the signal energy. Thus, ACF is energy characteristic of the signal. As one would expect, the signal and its copy are completely correlated (interconnected).

3. From a comparison of (2.58) and (2.59) it follows that the ACF is even function argument, i.e.

.

An important characteristic of the signal is correlation interval. The correlation interval is understood as the time interval, when shifted by which the signal and its copy become uncorrelated.

Mathematically, the correlation interval is determined by the following expression

,

or since is an even function

. (2.61)

In Fig. Figure 2.10 shows the ACF of an arbitrary waveform. If you build a rectangle with an area equal to the area under the curve for positive values ​​(the right branch of the curve), one side of which is equal to , then the second side will correspond to .

Digital signal delay circuits are required to temporarily O coordination of signal propagation along various paths digital device. Temporary mismatches between signals passing given paths can lead to critical timing races that disrupt the operation of devices. The transit time is affected by the parameters of the elements through which the transmission is transmitted. digital signals. By changing these parameters, you can change the signal propagation time. To change the delay time, electromagnetic delay lines, chains of logical elements, R.C.-chains. Using such elements, it is possible to obtain narrowing, widening of signals, narrowing with a shift relative to the front of the input pulse, etc.

Rice. 12.8. Short pulse former with a delay relative to the leading edge (a) and timing diagram (b)

Each logic element creates a time delay, so when input signal change in output signal level after the first logic element U 1 happens over time t health Similarly, after a time delay interval, the output signals of other inverters change ( U 2 ,U 3). The change in the state of the fourth element must be analyzed taking into account the fact that the inputs here are separate. Before the input signal arrives at the upper input of the logic element DD 4 was logical 1, and at the lower input was logical 0. Therefore, in steady state, the output of the circuit was high potential (logical 1).

After the input signal appears at the lower input of the element DD 4 is set to a logical one, the top one is also still 1. Therefore, at the output of the circuit after a while t zd.r will be set to logical 0. After passing through three logical elements, the input signal will change its value U 3 from 1 to 0 (this is the top input of the element DD 4). Output voltage of the circuit taking into account t z.r in element DD 4 will again become equal to 1. Consequently, the circuit generates a short pulse of duration 3 from the leading edge of the input signal t z.r with a shift relative to the leading edge by t health The falling edge of the input signal does not cause a change in the state of the circuit at the output, since by the time 1 appears at the upper input of the element DD 4 there is already a 0 at the bottom. Therefore, a 1 at the output is maintained until the next input pulse appears. The ongoing processes without taking into account the duration of the pulse fronts are presented on the time diagram (Fig. 12.8, b). The signal generated by the circuit has low level.

If the conjunctor DD 4 in the diagram (Fig. 12.8, A) is replaced with a disjunctor, and the number of inverters is made even, then the circuit will expand the input pulses for a time interval equal to nt z.r., where n– number of inverters in the delay circuit. The pulse expander circuit and timing diagram of its operation are shown in Fig. 12.9.

Rice. 12.9. Pulse expander circuit ( A) and timing diagram ( b)

From the timing diagram it is clear that the duration of the output pulse is longer than the duration of the input pulse by 4 t health

Only a few circuits of sequential pulse shapers are briefly considered. More information can be found in .

Monostibrators

Monotibrators (waiting multivibrators) belong to the group of regenerative circuits. This class of pulse devices generates time intervals of a given duration from an input trigger pulse of indefinite (but fairly short) duration (no longer than the duration of the generated pulse). To implement a standby multivibrator, a device with a transmission coefficient greater than one must be covered by regenerative (positive) feedback.

One of the possible single-vibrator circuits is shown in Fig. 12.10, A. The one-shot device is built on two logic elements of the 2I-NOT type by introducing a positive feedback(the output of the second element is connected to the input of the first).

IN original condition at the element output DD 2 there is level 1, and the output of the element DD 1 is a logical 0, since both its inputs have a 1 (the trigger pulses represent a negative voltage drop). When a triggering negative voltage drop is received at the input, level 1 will appear at the output of the first element. Positive drop across the capacitance WITH will arrive at the input of the second element. In this case, capacitance C will begin to charge through resistor R. Element DD 2 inverts this signal, and level 0 through the feedback circuit is supplied to the second input of the element DD 1. At the output of the element DD 2 level 0 is maintained as long as the voltage drop across the resistor R will not decrease to the value U pores during capacitor charging WITH(Fig. 12.10, b). The duration of the monostable output pulse can be determined using the expression

Rice. 12.10. Single-shot circuit ( A) and timing diagram ( b)

t and = C (R + R out) ln(U 1 /U por),

Where R out – output resistance of the first element; U pore – threshold voltage of the logic element.

The considered circuit can be implemented both on TTL microcircuits and on CMOS structures. However, the specifics of each type of logic imposes its own conditions. To build monovibrators, you can use flip-flops that have additional inputs S a and R and to force them to be set to one and zero states.

Single vibrators are produced in the form of independent microcircuits. The TTL series includes several waiting and controlled multivibrator microcircuits. The advantage of single-vibrators in microcircuit design is a smaller number of attached parts, greater temporal stability and broader functionality. Such microcircuits include monovibrators K155AG1 and K155AG3, as part of the CMOS series - 564AG1, 1561AG1. The operation of such microcircuits is described in detail in the literature.

Counters can be used to receive pulses of a given duration. Digital monostables are built on the basis of counters. They are used when the time interval must be very large or high demands are placed on the stability of the generated interval. In this case, the minimum duration obtained is limited only by the performance of the elements used, and the maximum duration can be any (unlike schemes using R.C.-chains).

The operating principle of a digital monostable is based on turning on the trigger by an input signal and turning it off after a time interval determined by the meter conversion factor. In Fig. Figure 12.11 shows an example of a circuit for obtaining a pulse of a given duration using a counter.

The operation of a single vibrator is illustrated by the diagrams in Fig. 12.11, b. In the initial state the trigger DD 2 on the inverse output has high level, which is at the entrance R sets the counter DD 1 to zero state. After the arrival of the input (triggering) pulse U in = 1 at moment t 1 trigger is set to single state. At its inverse output, a low level will be established, which will allow the programmable counter to count pulses DD 1. Counting pulses from the generator G continues to the value that is set by the programming inputs. After counting the specified number of pulses, a high-level signal is generated at the counter output UCT(moment t 2) which will return a trigger DD 2 to zero state. In this case, the inverse output of the trigger will again be set to a high level, and the counter will return to its original state.

Rice. 12.11. Block diagram ( A) and timing diagrams

(b) digital monostable

A common disadvantage of such schemes is the random error associated with the randomness of the phase of the master oscillator at the time of startup. The error can be up to a period of the clock frequency and decreases with increasing generator frequency. This drawback can be eliminated by schemes with controlled start of the generator (the generator turns on when a trigger pulse appears).

The use of counters with a programmable division coefficient as part of a one-shot device makes it possible to obtain a pulse of any duration. The 564IE15 chip, for example, consists of five subtractive counters, the conversion modules of which are programmed by parallel loading of data in binary code. Higher stability of the output pulse duration is ensured by the use of crystal oscillator clock frequency.

Can an impulse tell you anything? - you say. An impulse is just that, an impulse, only rectangular in shape.

But the fact of the matter is that until now we have only observed similar pulses on the oscilloscope screen, say, during the setup of an electronic switch, and by their presence we judged the serviceability of the generator. If you use a rectangular pulse as a control signal and apply it, for example, to the input of an AF amplifier, then from the shape of the output signal you can immediately evaluate the operation of the amplifier and name its shortcomings - low bandwidth, insufficient gain at low or higher frequencies, self-excitation in some frequency range.

Take a broadband voltage divider, used, for example, in homemade measuring instruments or oscilloscopes. A rectangular pulse “passed” through it will tell you the exact parameters of the parts necessary to obtain a constant signal division coefficient in a wide frequency range.

To make this clear, let's first get acquainted with some parameters of the pulse signal, which are often mentioned in descriptions of various generators, automation devices and computer technology. For example in Fig. 97 shown" appearance"a slightly distorted (compared to rectangular) pulse, so that its individual parts are more clearly visible.

One of the pulse parameters is its amplitude (Umax), the highest pulse height without taking into account small emissions. The duration of the rise of the pulse characterizes the duration of the front tf, and the duration of the decrease is characterized by the duration of the decline tс. The duration of the “life” of the pulse is determined by the duration ti - the time between the beginning and end of the pulse, usually counted at the level of 0.5 amplitude (sometimes at the level of 0.7).

The top of the impulse can be flat, with a collapse or a rise. A rectangular pulse has a flat top, and the rise and fall are so steep that it is impossible to determine their duration using an oscilloscope.

The pulse signal is also evaluated by its duty cycle, which shows the relationship between the pulse duration and the pulse repetition period. Duty duty is the quotient of dividing the period, not the duration. In the one shown in Fig. 97, in the example, the duty cycle is 3.

Now, after a brief introduction to the pulse and its parameters, we will build a rectangular pulse generator necessary for subsequent experiments. It can be made both on transistors and on microcircuits. The main thing is that the generator produces pulses with steep rises and falls, as well as with the flattest possible top. In addition, for our purposes, the duty cycle should be within 2-3, and the pulse repetition rate should be approximately 50 Hz in one mode, and 1500 Hz in the other. You will find out later what causes the frequency requirements.

The easiest way to meet the requirements is a generator based on a microcircuit and a transistor (Fig. 98). It contains few parts, is operational when the supply voltage is reduced to 2.5 V (in this case, the signal amplitude mainly drops) and allows you to obtain output pulses with an amplitude of up to 2.5 V (at the specified supply voltage) with a duty cycle of 2.5.

Actually, the generator itself is made using elements DD1.1 - DD1.3 according to the well-known multivibrator circuit. The pulse repetition rate depends on the resistance of resistor R1 and the capacitance of the capacitor currently connected by switch SA1. In the position of the moving contact of the switch shown in the diagram, capacitor C1 is connected to the generator, so the pulses at the output of the generator (pin 8 of element DD1.3) follow with a frequency of 50 Hz (following period 20 ms). When the moving contact of the switch is placed in the lower position according to the circuit, capacitor C2 is connected and the repetition frequency becomes approximately 2000 Hz (repetition period 0.5 ms).

Next, the pulse signal is supplied through resistor R2 to the emitter follower, made on transistor VT1. From the motor of the variable resistor R3, which is the load of the repeater, the signal is supplied to the output terminal XT1. As a result, rectangular pulses with an amplitude from several tens of millivolts to several volts can be removed from the XT1 and XT2 terminals. If for some reason even the minimum signal turns out to be in excess (for example, when testing a very sensitive amplifier), the output signal can be reduced either by connecting resistor R3 between the upper terminal of the circuit and the emitter of the transistor with a constant resistor with a resistance of 1-3 kOhm, or by using an external divider voltage.

A few words about the details. The generator can operate NAND elements of other K155 series microcircuits (say, K155LA4), as well as any KT315 series transistor. Capacitor C1 - K50-6 or another, designed for a voltage of at least 10 V; C2 - any, possibly smaller in size. Resistors - MLT-0.125 and SP-1 (R3), power source - battery 3336. The generator consumes less than 15 mA, so such a source will last for a long time.

Since there are few parts in the generator, there is no need to provide a drawing printed circuit board- develop it yourself. Mount the board with the parts and the power supply inside the case (Fig. 99), and on its front wall place the range switch, power switch, variable resistor and clamps.

The next stage is checking and setting up the generator using our oscilloscope. Connect the oscilloscope input probe to pin 8 of the microcircuit, and the ground probe to the common wire (XT2 terminal). The oscilloscope is still working automatic mode(the "AUTO-STANDBY" button is released), synchronization is internal, input is open to eliminate distortion of the signal following at a low frequency). The oscilloscope's input attenuator can set the sensitivity to, say, 1 V/div, and the sweep duration switches can set the sweep duration to 5 ms/div.

After applying power to the generator and setting switch SA1 to the position shown in the diagram, an image in the form of two parallel images will appear on the oscilloscope screen.

line lines (Fig. 100, a), composed of moving “strokes”. This is what an unsynchronized image of a pulse signal looks like.

It is enough now to switch the oscilloscope to standby mode (press the "AUTO - STANDBY" button) and set synchronization from a positive signal by turning the "SYNC." knob. to the extreme clockwise position so that the image on the screen “stops” (Fig. 100, b). If the image jitters a little, use the sweep length adjustment knob to get it in better sync.

Determine the duration of the pulse repetition period and, if necessary. set it to 20 ms by selecting resistor R1.

It is difficult to accurately measure the period with a set sweep duration, so use a simple technique. For this trigger, set the sweep duration to 2 ms/div. More than one should appear on the screen stretched image pulse (Fig. 100, c), the length of the top of which will be approximately 3.5 divisions, i.e., the pulse duration will be equal to 7 ms.

Then, at the same sweep duration, set the synchronization with a negative signal by turning the "SYNC." knob. to the extreme counterclockwise position. You will see an image of a pause on the screen (Fig. 100, d), since the oscilloscope sweep is now triggered by the decay of the pulse. The line length is 6.5 divisions, which means the pause duration is 13ms. The sum of the pulse and pause durations will be the value of the pulse repetition period (20 ms).

Similarly, check the operation of the generator on the second range by setting the moving contact of the switch to the lowest position according to the diagram (“2 kHz”). In this case, set the oscilloscope sweep duration to, for example, 0.1 ms/div. The pulse repetition period in this range should be 0.5 ms, which corresponds to a repetition rate of 2000 Hz. There is no need to adjust anything in the generator, since frequency accuracy in this range does not play a special role. In case of a significant deviation of the frequency from the specified one, it can be changed by selecting capacitor C2.

After this, switch the input probe of the oscilloscope to the XT1 terminal and check the operation of the output signal amplitude regulator - variable resistor R3. You will probably notice that when the variable resistor is installed in the upper position in the circuit, the maximum amplitude of the pulses will be slightly less than on a multivibrator. This is explained by the action of the emitter follower, the transmission coefficient of which is less than unity due to the drop in part of the signal at the emitter junction of the transistor.

The generator is ready, you can conduct experiments. Let's start by checking the impulse action of simple RC circuits: differentiating and integrating. First, connect a differentiating circuit consisting of a capacitor and a variable resistor to the generator output (Fig. 101). Place the resistor slider in the lowest position according to the diagram, and set the range on the generator to “50 Hz” and the maximum amplitude of the output signal. In this case, on the oscilloscope screen (it operates in standby mode with synchronization from a positive signal, sweep duration is 5 ms/div., sensitivity is 1 V/div.) you will see an image of pulses with a beveled top (Fig. 102, a). It is easy to notice that the impulse seemed to fall along the decline line, which is why the scope of the image increased.

The pulse distortion will increase and the image scope will increase as the variable resistor slider moves up the circuit. Already with a resistor resistance of about 4 kOhm, the swing will almost reach twice the pulse amplitude

(Fig. 102, b), and with a further decrease in resistance (to 1 kOhm), only pointed peaks will remain from the pulse at the site of the front and decline. In other words, as a result of differentiation from a rectangular pulse it will be possible to obtain two pointed ones - positive (along the front) and negative (along the fall).

In addition, differentiation allows you to “shorten” the pulse in time - after all, the pulse duration is measured at the level of 0.5 of its amplitude, and at this level the pulse width changes smoothly when the variable resistor knob is turned).

The differentiating properties of the circuit depend on the pulse repetition rate. It is enough to move the generator range switch to the “2 kHz” position - and the bevel of the top will practically disappear. Pulses following at such a frequency are passed through by our differentiating chain with virtually no distortion. To get the same effect as in the previous case, the capacitance of the capacitor must be reduced to 0.01 µF.

Now swap the parts (Fig. 103) - you get an integrating chain. Place the variable resistor slider in the leftmost position according to the diagram, i.e., output the resistor resistance. The signal image will remain almost the same as at the output of the generator before connecting the chain. True, the decay of the pulses will become slightly curved - the result of the discharge of the capacitor, which has time to charge during the pulse.

Begin to smoothly move the resistor slider to the right according to the diagram, i.e., enter the resistance of the resistor. Immediately the front of the pulse and the fall will begin to round off (Fig. 104, c), and the amplitude of the signal will fall. At maximum resistor resistance, the observed signal sounds like a sawtooth (Fig. 104,b).

What is the essence of integration? From the moment the front of the pulse appears, the capacitor begins to charge, and at the end of the pulse, it begins to discharge. If the resistance of the resistor or the capacitance of the capacitor is small, the capacitor manages to charge up to the amplitude value of the signal and then only the front and part of the top of the pulse “collapses” (Fig. 104, a). In this case, we can say that the time constant of the integrating circuit (the product of capacitance and resistance) is less than the pulse duration. If the time constant is comparable to or exceeds the duration of the pulse, the capacitor does not have time to charge completely during the pulse and then the amplitude of the signal on it drops (Fig. 104, b). Of course, the nature of integration depends not only on the duration of the pulses, but also on the frequency of their repetition.

To verify this, again output the resistor, set the generator to “2 kHz” and change the oscilloscope sweep duration accordingly. A picture of already integrated impulses will appear on the screen (Fig. 104, c). This is the result of the "interaction" of the emitter follower resistance and the capacitor capacitance. Introduce at least a small resistance with a variable resistor - and you will see a triangular-shaped signal on the oscilloscope screen (Fig. 104, d). Its amplitude is small, so you will have to increase the sensitivity of the oscilloscope. Isn’t it true that the linearity of the process of charging and discharging the capacitor is clearly visible?

In this example, the time constant of the integrating circuit is slightly longer than the pulse duration, so the capacitor has time to charge only to a very small voltage.

It's time to talk about the practical use of rectangular pulses, for example, to evaluate the performance of an audio amplifier. True, such a method is suitable for a kind of express analysis and does not provide a comprehensive picture of the amplitude-frequency characteristics of the amplifier. But it allows you to objectively evaluate the amplifier’s ability to transmit signals of certain frequencies, resistance to self-excitation, as well as the correct choice of parts between cascade connections.

The principle of testing is simple: first, rectangular pulses with a repetition rate of 50 Hz and then 2000 Hz are applied to the input of the amplifier, and the shape of the output signal is observed at the load equivalent. By the distortion of the front: the top or the bottom, the characteristics of the amplifier and its stability of operation are judged.

For example, you can examine an AF amplifier with a tone block (or another broadband amplifier). It is connected to a generator and an oscilloscope according to Fig. 105. The generator range switch is set to the “50 Hz” position, and the output signal is such that, with maximum amplifier gain and approximately average positions of the tone control knobs, the signal amplitude at the equivalent load corresponds to the rated output power, for example 1.4 V (for a power of 0.2 W at a load resistance of 10 Ohms). The picture on the screen of an oscilloscope connected to a load equivalent may correspond to that shown in Fig. 106, a, which will indicate insufficient capacitance of the separating capacitors between the amplifier stages or the capacitor at the output of the amplifier - a load is connected through it.

To verify, say, the last assumption, it is enough to move the input probe of the oscilloscope directly to the output of the amplifier - before the coupling capacitor. If the bevel of the top decreases (Fig. 106, b), then the conclusion is correct and for better reproduction of lower frequencies, the capacitor capacity should be increased.

Similarly, they look at the images of the pulses before and after the coupling capacitors between the amplifier stages and detect the one whose capacitance is insufficient. If the amplifier does not transmit low frequencies well at all, narrow peaks may be observed on the oscilloscope screen at the place of the rise and fall of the pulses, as was the case with strong differentiation. But a more complete picture of the state of the amplifier is obtained when pulses with a frequency of 2000 Hz are applied to its input. It is believed that the front and fall reflect the passage of higher frequencies of the sound range, and the top - the lower ones.

If everything is in order in the amplifier and it uniformly transmits the signal over a wide frequency band, then the output pulse (signal at the load equivalent) will correspond in shape to the input pulse (Fig. 107, a). In the case of a “blockage” of the front and decline (Fig. 107, b), we can assume that the gain at higher frequencies has decreased. An even greater reduction in gain at these frequencies will be recorded in the image shown in Fig. 107, a.

Many other options are possible: the gain drops by lower frequencies(Fig. 107, d), a slight increase in gain at lower frequencies (Fig. 107, e), a drop in gain at low and medium (dip at the top) frequencies (Fig. 107, f), a small time constant of interstage connections (Fig. 107, g) - usually the capacitance of the transition capacitors is small, the gain increases at lower (Fig. 107, h) or higher (Fig. 107, i) frequencies, and the gain decreases in a certain narrow range (Fig. 107, j).

And here are two examples of images of the output pulse (Fig. 107, l, m), when the amplifier has resonant circuits.

Almost most of these images can be observed by changing the positions of the tone control knobs for lower and higher frequencies. At the same time as viewing the images, it would be a good idea to take the amplitude-frequency response of the amplifier and compare it with the “readings” of the pulses.

And about one more example of using rectangular pulses - to configure broadband voltage dividers. Such a divider, for example, is in our oscilloscope; it can be in a voltmeter or millivoltmeter of alternating current. Since the frequency band of the measured signals can be very wide (from units to millions of hertz), the divider must pass these signals with the same attenuation, otherwise measurement errors are inevitable.

You can, of course, check the operation of the divider by reading its amplitude-frequency characteristic, which will tell you in which direction the value of a particular element should be changed. But this matter is much more labor-intensive compared to the rectangular pulse analysis method.

Take a look at fig. 108, a - it shows a diagram of a broadband compensated voltage divider. If at lower frequencies it would be possible to get by only with resistors, the resistance of which determines the transmission coefficient (or division coefficient) of the divider, then at higher frequencies, in addition to resistors, capacitors in the form of installation capacitance, input capacitance, and capacitance of connecting conductors participate in the operation of the divider. Therefore, the gain of the divider at these frequencies can change significantly.

To prevent this from happening, the divider uses capacitors that shunt resistors and make it possible to compensate for possible changes in the transmission coefficient at higher frequencies. Moreover, capacitor C2 can be an installation capacitance, sometimes reaching tens of picofarads. Resistor R2 can be the input resistance of a device (oscilloscope or voltmeter).

The divider will become compensated if a very specific ratio of the resistances and capacitances of the divider is ensured, which means that the transfer coefficient of the divider will be uniform regardless of the frequency of the input signal. For example, if a divisor by 2 is used, then the condition R1* C1=R2*C2 must be met. With other ratios, the uniformity of signal transmission of different frequencies will be disrupted.

The principle of testing a compensated divider using rectangular pulses is similar to the principle of testing an amplifier - by applying a signal with a frequency of 2000 Hz to the input of the divider, its shape at the output is observed. If the divider is compensated, the shape (but, of course, not the amplitude) of the signals will be the same. Otherwise, the front and fall will be “overwhelmed” or the top will be distorted - evidence of uneven transmission of signals of different frequencies by the divider.

If, for example, the signal image is as shown in Fig. 108, b, it means that at higher frequencies the divider transmission coefficient drops due to high resistance at these frequencies the chains are R1C1. The capacitance of capacitor C1 should be increased. In the event of pulse distortion shown in Fig. 108, in, it is necessary, on the contrary, to reduce the capacitance of capacitor C1.

Try to independently create dividers with different division coefficients (for example, 2, 5, 10) from high-resistance resistors (100...500 kOhm) and capacitors of different capacities (from 20 to 200 pF) and achieve full compensation by selecting capacitors.

In this work, you will notice the influence of the oscilloscope itself on the measurement results - after all, its input capacitance is tens of picofarads, and

input impedance is about megaohm. Remember that the oscilloscope has a similar effect on all high-impedance circuits, as well as frequency-dependent ones. And this sometimes leads either to erroneous results, or even makes it impossible to use an oscilloscope, say, to analyze the operation and measure the frequency of radio frequency generators. Therefore, in such cases, you should use an active probe - an attachment to the oscilloscope, which allows you to maintain its high input impedance and reduce the input capacitance tens of times. A description of such an attachment will be published in the next issue of the magazine.

Now that you have become familiar with the ability of a rectangular pulse to prompt a “diagnosis” and control “treatment,” let’s assemble another attachment. This is a voltage divider, with the help of which an oscilloscope will be able to monitor circuits with voltages up to 600V, for example, in television receivers (as you know, the OML-2M oscilloscope allows voltages of up to 300V to be supplied to the input).

The divider is formed by only two parts (Fig. 109), which make up the upper arm of the previous diagram. The lower arm is concentrated in the oscilloscope itself - this is its input resistance and total input capacitance, including the capacitance of the remote cable with probes.

Since you only need to halve the input signal, resistor R1 must have the same resistance as the input resistance of the oscilloscope, and the capacitance of capacitor C1 must correspond to the total input capacitance of the oscilloscope.

The divider can be made in the form of an adapter with an XP1 probe at one end and an XS1 socket at the other. Resistor R1 must have a power of at least 0.5 W, and the capacitor with a rated voltage of at least 400 V.

Setting up the divider is greatly simplified by using our pulse generator. Its signal is fed to the XP1 socket of the divider and the ground probe of the oscilloscope. First, set the range to “50 Hz” on the generator, turn on the standby mode on the oscilloscope and open entrance. Touch the input probe of the oscilloscope to the probe XP1 of the divider (or clamp XT1 of the generator). By selecting the sensitivity of the oscilloscope and the amplitude of the generator output signal, the range is achieved

image equal to, say, four divisions.

Then switch the input probe of the oscilloscope to socket XS1 of the divider. The image size should be reduced exactly by half. More accurately, the divider transmission coefficient can be set by selecting the resistor R1 of the divider.

After this, set the “2 kHz” range on the generator and select the capacitor C1 (if necessary) to achieve the correct pulse shape - the same as at the input of the divider.

When using such a divider to check the operating modes of television scanning units using the signal images given in the instructions and various articles, the sensitivity of the oscilloscope is set to 50 V/div, and the test is carried out at closed entrance oscilloscope. As before, the countdown is carried out on a grid scale, but the results are doubled.

DESCRIPTION

INVENTIONS

Union of Soviets

Socialist

State Committee

USSR for Inventions and Discoveries

A.V. Kozlov (71) Applicant (54) RECTANGULAR PULSE DELAY DEVICE

The invention relates to measuring. body and computer technology and can be used, in particular, in extreme correlation systems for determining the speed of movement, in correlation flow meters, in pulsed automation devices.

A pulse delay device is known, containing a pulse generator, an input control trigger, an AND element, a controlled frequency divider (1 j.

The disadvantage of the device is that when the pulses are delayed, their duration is not preserved.

A pulse delay device is also known, containing a pulse generator, three AND elements, two control triggers, a reverse counter, a controlled frequency divider, and a zero f 2 decoder.

However, the device has a rather complex control circuit due to the use of a reversible counter.

The closest in technical essence to the proposed one is a rectangular pulse delay device containing a pulse generator, a delay time register, a controlled frequency divider consisting of a binary counter, a reset and write circuit and two AND elements, 5 the first and second inputs of which are connected respectively, with the outputs of the delay time register and the first output of the reset and set circuit, and the outputs of the elements are connected to the setting S-inputs of the counter, the first and second elements are AND and RS flip-flops, a binary counter and a comparison circuit, the output of which is connected to the reset inputs of the RS flip-flops , and its inputs are connected to the information outputs of a binary counter and a controlled frequency divider, the output of which is connected to the setting input of the second

An RS flip-flop, the output of which is connected to the input of the reset and write circuit and is the output of the device, the pulse generator, through the first inputs of the AND elements, is connected to the control inputs of a binary counter and a controlled frequency divider, respectively, the reset inputs of which are connected to the second output of the reset and write circuit, the input signal source is connected to the second input of the second AND element and to the setting input of the first R5, -flip-flop, the output of which is connected to the second input of the first AND element (3).

The disadvantage of the device is that it does not provide a delay of the input pulse in the case where the time between the end of the previous input pulse and the beginning of the next pulse is less than the delay time, since in this condition the device has not yet generated the delayed previous pulse and therefore cannot accept the next input pulse . Indeed, if the formation of the previous delayed pulse has not been completed, then when the next pulse arrives at the input of the device, it will not change the state of the first WB trigger, since the latter is already in the “1” state, but will open the second AND element. At the same time. The binary counter will receive from the generator a number of pulses proportional to the duration of this input pulse. The binary counter code will become proportional to the sum of the durations of the previous and subsequent 75 input pulses, i.e. the duration of the generated output pulse will be equal to the total duration, which is a malfunction of the delay device. The problem of delaying pulses with variable duration under the condition described above arises in extreme correlation velocity measurement systems, in correlation flow meters and other pulse devices. These devices are synchronized with a tunable clock frequency.

In each cycle, only one rectangular pulse is formed, the duration of which is determined by the measured parameter in this cycle. This impulse must be delayed for a period of one tkt. In this case, the leading edge of the pulse coincides with the beginning of the clock cycle, therefore, in order to delay the pulse by 45 clock cycles, it is necessary and sufficient to delay only the trailing edge of the pulse, since its leading edge is associated with the beginning of the clock cycle and is determined by the clock frequency pulse. Time between 50 two rectangular pulses. in such named devices there is always less than a delay time equal to the clock frequency change, therefore the task is to improve the considered rectangular pulse delay device to fulfill the specified requirement °

The purpose of the invention is to expand the functionality of the 6O device for delaying rectangular pulses.

This goal is achieved by the fact that in a rectangular pulse delay device containing a pulse generator, a controlled frequency divider g5, two AND elements, two RS flip-flops, a delay time register, the output of which is connected to the information input of the controlled frequency divider, the output of the pulse generator is connected with the first inputs of the AND elements, the output of the first RS trigger is connected to the second input of the first AND element, the output of which is connected to the control input of the controlled frequency divider, and the output of the second RS trigger is the output of the device, a switch, a shaper, the input of which is the input of the device, are introduced, and the output of the shaper is connected to the input of the commutator, the third RS trigger, the output of which is connected to the second input of the second AND element, the OR element, the output of which is connected to the R input of the second RS trigger, the second and third controlled frequency dividers, the information inputs of which are connected to output of the delay time register, the outputs of the first and second controlled frequency dividers are connected to the inputs of the element

HJIH ooT eT T e o K R-inputs of the first and third RS-flip-flops, the S-inputs of which are connected to the corresponding outputs of the switch, the output of the pulse generator is connected to the control input of the third controlled frequency divider, the output of which is connected to the control input of the switch and

The S-input of the second R 5 flip-flop, the output of the second AND element is connected to the control input of the second controlled frequency divider.

Indeed, the introduction of new elements and new connections makes it possible to delay rectangular pulses for a time equal to the period of the tunable clock frequency, while the time between two delayed pulses is less than the delay time.

To eliminate the influence of the subsequent pulse on the formation of the delayed previous pulse, a commutator, two RS flip-flops, two AND elements, and two controlled frequency dividers are used. The switch connects one or the other in turn at each cycle of operation of the device.

An RS trigger, therefore, a short pulse corresponding to the falling edge of the delayed pulse is supplied from the output of the shaper in turn to the indicated RS flip-flops, and the pulses are delayed in turn at the first and second controlled frequency dividers. This eliminates the influence of the subsequent input pulse on the generation of the previous delayed pulse and allows the subsequent pulse to be delayed.

In fig. 1 is given block diagram the proposed delay device for rectangular pulses; on

1003321 fig. 2 - timing diagrams explaining the operation of the delay device.

The device contains a driver

1, switch 2, pulse generator

3, R5 flip-flops 4 and 5, AND elements 6 and 7, controlled frequency dividers 8-10 - 5, delay time register 11, OR element 12, output RS flip-flop 13.

The input of shaper 1 is the input of the device, and its output is connected to the input of switch 2, the output of which is connected to the S-inputs of R5 flip-flops 4 and 5, the output of pulse generator 3 is connected to the control input of the controlled divider

8 frequencies and first element inputs 15

And b and 7, the outputs of which are connected respectively to the control inputs of controlled frequency dividers 9 and

10, the outputs of which are connected respectively to the R inputs of the R5 flip-flops

4 and 5 and with the inputs of the OR element, the output of which is connected to the R-input

RS trigger 13, the output of the delay time register 11 is connected to the information inputs of the controlled frequency dividers 8-10, the output of the controlled frequency divider 8 is connected to the controlled input of the switch 2 and to

5-input RS flip-flop 13, the output of which is the output of the delay device.

Shaper 1 is designed to generate a short pulse, which corresponds to the falling edge of the input delayed pulse, Ç5 arriving at its input. Switch 2 in turn connects the output of driver 1 to the S inputs of RS flip-flops 4 and 5. Pulses from generator 3, passing through divider 8, form 40 clock pulses, the period of which is equal to the delay time and is determined by register code 11. Clock pulses are supplied to the control input of the switch and S-input45

RS-trigger 13, which ensures switching of pulses from the output of the shaper with a frequency equal to the clock frequency, and the formation of the leading edge of the delayed pulse íà Output-50 of RS-trigger 13 according to the clock frequency pulse, i.e. from the beginning of the next measure. Dividers 9 and 10 form a pulse delayed by a period of the clock frequency, OR element 12 carries out the operation of combining the outputs of dividers 9 and 10, so each delayed pulse arrives from the outputs of dividers 9 and 10 íà R-âõñä

RS trigger 13, and the trailing edge of a delayed 60 pulse is formed at its output.

The device works as follows.

The output clock pulses generated at the output of divider g5 synchronize the operation of not only the delay device, but also the entire device in which this device is used. The input of delay device 1 receives rectangular pulses that must be delayed for the duration of one clock cycle. The leading edges of all pulses coincide with the beginning of the clock cycles, so the clock frequency pulses are supplied to the 5-input RS of flip-flop 13, while delayed pulses are formed at its output, the leading edges of which coincide with the beginning of the clock cycles. Pulses from the output of shaper 1, passing through switch 2, alternately, through a clock, arrive at the S-inputs of flip-flops 4 and 5.

With the arrival of such a pulse, rectangular pulses are formed on these flip-flops (alternately at each clock cycle) using the AND element 6 or 7 and the divider 9 or 10, the duration of which is equal to the clock frequency period, since the division coefficients of the dividers 8-10 are equal and are determined by the register code. 11 delay time. The falling edges of these pulses coincide with the output short pulses of dividers 9 and 10, since these short pulses arrive at the R-inputs of RS flip-flops 4 and 5 and set a “0” signal at their outputs, stopping the passage of pulses from generator 3 in turn in each tact through the elements

And b or 7 to the inputs of dividers 9 or

10. Pulses from the outputs of dividers and 10, passing through the OR element, are summed up and fed to the R-input RQ— of trigger 13, which before the arrival of these pulses in each clock cycle is in the “1” state. Incoming pulses from the R-input transfer this trigger to the “.0” state, forming the trailing edge of delayed pulses. Thus, at the output of RS trigger 13 a sequence of rectangular pulses is formed, delayed by one clock cycle compared to the sequence of input pulses.

The proposed square wave delay device extends the functionality of the prototype by providing pulse delay provided that the time between two input pulses is less than the required delay time, which can be changed by changing the delay time register code. It can be used in correlation meters of speed, flow and other similar pulse devices ° At the same time clock frequency and a pulse generator are used to synchronize the operation of the entire meter. In addition, the delay circuit is significantly simplified, since the operations of measuring, storing and restoring the duration of the delay are eliminated1003321

Formula of invention

ro input pulse. Cost reduction when using the proposed device in the above-mentioned meters depends on the required accuracy and discreteness of time changes, delay, determined by the number of digits of controlled frequency dividers. In the prototype, this requirement affects the number of bits of the binary counter, in which the duration of the delayed pulse is recorded. This counter with a duration measurement circuit is absent in the proposed device, which could be replaced by two prototype circuits with additional elements in the mentioned meters. Using this device instead of two prototype circuits, the number of chips can be reduced, which reduces costs. (The pulse delay error is also halved, since only the falling edge of the pulse is delayed, and the leading edge coincides with the clock pulses, so the pulse delay error is determined only by the falling edge delay error.

A rectangular pulse delay device containing a pulse generator, a controlled frequency divider, two AND elements, two RS flip-flops, and a time register. delay, the output of which is connected to the information input of the controlled frequency divider, the output of the pulse generator is connected to the first inputs of the AND elements, the output of the first RS trigger is connected to the 40 second input of the first AND element, the output of which is connected to the control input of the controlled frequency divider, and the output of the second k5 -trigger is the output of the device, differing in that, in order to expand the functionality of the device, a switch, a driver, the input of which is the input of the device, and the output of the driver is connected to the input of the switch, are introduced into it , the third g5 flip-flop, the output of which is connected to the second input of the second AND element, the OR element, the output of which is connected to

By the A-input of the second R5-trigger, the second and third controlled frequency dividers, the information inputs of which are connected to the output of the delay time register, the outputs of the first and second controlled frequency dividers are connected to the inputs of the OR element and, accordingly, to the R-inputs of the first and third K3-triggers, 5-inputs of which are connected to the corresponding outputs of the switch, the output of the pulse generator is connected to the control input of the third controlled frequency divider, the output of which is connected to the control input of the switch and the 5-input of the second 95-flip-flop, the output of the second element And is connected to the control input of the second controlled frequency divider.

Sources of information taken into account during the examination

R 308499, class. N 03 K 5/1 3, 1969.

R 396822, class. N 03 K 5/153, 1971.

R 479234, class. N 03 K 5/153, 1973 (prototype).

VNIIPI Order 1588 44 Tier 934 Subscription

Branch of PPP "Patent", Uzhgorod, Proektnaya St., 4

S. Andrianov

Digital ICs are widely used in the development and creation of many pulsed devices, since it does not require the calculation of transistor switches, and there is no need to coordinate signal voltage levels when these devices operate with the same type of logic.

Let's look at some of these devices based on digital ICs. When analyzing their work, everything p-n transitions will be considered ideal switches with a threshold voltage Uo.

Let's start with the pulse edge delay device, which is the basis of all devices discussed below. Using his example, moreover, it is easiest to understand the features of the operation of pulsed devices on digital ICs.

The device diagrams are shown in Fig. 1, and diagrams of voltages and currents in its various circuits are in Fig. 2 (hereinafter, examples of devices are given in relation to DTL microcircuits of the K217 series, which does not limit the generality of the conclusions in relation to TTL microcircuits). In the initial state, the input of the device (Fig. 1, b) a logical 0 signal is supplied, i.e. current i 0 is diverted to the common wire through public key previous element. Capacitor C1 charged to voltage Uo open diode VI. At a moment in time t x (Fig. 2) a logical one signal comes to the input, which is equivalent to disconnecting the device input from the common wire. Diodes VI, V3 close and disconnect the signal source from the device input.

Now the current I 0 charges the capacitor C1 to tension 2 U 0 . In this case, the voltage at the point b becomes equal to 3U 0 . Diodes open V4, V5 and transistor V6 - an inverted delayed edge of the input pulse appears at the device output.

When the cut passes, the device input will again close to the common wire, diodes V2, V4 And V5 will close and the capacitor C1 for very short time will discharge through the diode VI to tension Uo. Transistor V6 will close and the device will return to its original state. In order for the edge delay of the input pulse to occur without inversion, there must be an inverter at the output of the device.

Rice. 1. Functional (A) and principled (b) pulse edge delay device circuits

Pulse cutoff delay device, the circuit and timing diagrams of which are shown in Fig. 3 differ from the pulse edge delay device only in that an inverted signal is supplied to its input. And since it is controlled by a positive voltage drop, there is a delay in the cutoff of the input pulse.

The next pulse device is a pulse delay device. It is essentially two edge delay stages. Having passed through the first stage, the pulse is inverted with an edge delay, while the second stage works exactly the same as in the previous device. As a result of the delay of the front and the cutoff for the same time, the pulse received at the input is delayed in time while maintaining its previous duration.

Rice. 2. Timing diagrams of voltages and currents in the circuits of the pulse edge delay device

Rice. 3. Pulse cutoff delay device:

a - functional diagram; b- voltage timing diagrams

These features determine the areas of use of the considered time delay devices. The second of them is best used when the pulse duration or duration ratio is unknown.

A pulse former of a given duration (Fig. 5) consists of a coincidence element D2 (2I-NOT), to one of the inputs of which the input pulse is supplied directly, and to the other - with an edge delay and with inversion. The output signal is a logical zero pulse, the duration of which is equal to the delay time of the front of the input pulse.

Rice. 4. Pulse time delay device:

A - functional diagram; 6 - voltage timing diagrams

Rice. 5. Device for generating pulses of a given duration: A - functional diagram; b - voltage time diagrams

Based on such a device, a frequency-voltage converter can be designed. To do this, it is enough to turn on an integrating circuit at its output. The principle of operation of the converter is that the constant component of a periodic pulse signal is inversely proportional to the duty cycle (the ratio of the period to the pulse duration), and, therefore, with a constant duration, it is directly proportional to the frequency. Constant component impulse voltage is distinguished by an integrating chain.

The next pulse device is a self-oscillating multivibrator, the circuit of which is shown in Fig. 6. It consists of two identical (symmetrical case) pulse formers of a given duration, assembled on elements Dl… Df, diodes VI, V2, and capacitors C1 and C2. Element D5 designed to start a multivibrator and establish a self-oscillating operating mode after turning on the power. The oscillation period is determined by the sum of the pulse durations generated in the arms of the multivibrator.

The device works as follows. After turning on the power, when the capacitors C1 And C2 are not yet charged, a logical one signal is observed at the outputs of the multivibrator arms. Element D5 generates a logical zero signal, i.e. closes the corresponding input of the element D1 to the common wire. Consequently, only the capacitor gets the opportunity to charge C2. From the moment the capacitor starts charging C2 and until the end of the impulse formation by the elements D2, D4 at the element output D4 and at the corresponding input of the element D1 a logic zero signal is supported, which prevents the capacitor C1 charge until the capacitor charging cycle is complete C2, and vice versa. As it is now at the entrances. element D5 signals of logical zero and one appear alternately in antiphase, then at the output of the element D5 a logical one signal is observed all the time and it has virtually no effect on the further operation of the device.

The waiting multivibrator is a combination of an edge delay device and an RS flip-flop, the state of which is changed by a logical zero (Fig. 7). Trigger pulses, which are logical zero signals, go to the input of the element D2. In the initial state, the output of this element is logical zero, and the output of the element D3 - unit. The trigger will remain in this state for as long as desired until the trigger pulse arrives.

Rice. 6. Functional diagram of a self-oscillating multivibrator

Rice. 7. Functional diagram of a waiting multivibrator

At the moment of launch, the trigger switches to another state, and from the output of the element D2 to the input of the edge delay device formed by the element D1, diode VI and capacitor C1, a logical one signal arrives. The delay device inverts the signal with a time delay, which ensures that the flip-flop switches back and restores its original state.

The waiting multivibrator considered here has two outputs: for logical zero pulses - the output of the element D3, for logical unit pulses - element output D2.

Calculating timing characteristics is not difficult. Analysis of transient processes in the device according to the diagram in Fig. 1, b for the edge delay time t hell gives the following expression:

Where U al - supply voltage.

When the ratio is small Uo/ U n 1 you can use the approximate formula

(2)

then at U 0 =0,7 V, U p1 =6 V, the relative error of the calculated delay time will be less than 6%, and with U 0 =0.7 V and U P1 = 5 V - less than 8%.

Temperature stabilization of the considered pulsed devices can be carried out by setting the appropriate temperature dependence of the bias supply voltages so as to compensate for the temperature drift of the threshold voltage p-n transitions. From expression (1), taking into account the temperature dependence only Uo And U nl, the expression for the temperature drift of the delay time is obtained:

By equating the value of the temperature drift of the delay time to zero and solving the resulting equation regarding the temperature drift of the voltage of the bias source, in the example under consideration (see Fig. 1, b) - U nU we obtain the required dependence of the supply voltage on temperature, ensuring stabilization of the delay time when the ambient temperature changes:

(4)

Rice. 8. Circuit of a bias voltage source (supply) with temperature dependence of the output voltage to compensate for thermal drift

Let us now consider the calculation of a voltage source with the required temperature dependence. For example, let's take a stabilizer made according to the diagram in Fig. 8. Here is a field effect transistor V4 - source of stable current. From the collector of the transistor V5 exemplary tension is removed. On a transistor V6 current amplifier assembled. Load Rn are parallel-connected bias circuits of logic elements that require stabilization of the bias voltage with a certain temperature dependence. In order for the temperature dependence of the output voltage to meet the necessary requirements, the following relation must be satisfied:

(5)

Suppose you need to stabilize the bias voltage of three logic elements of the K217 series with the temperature dependence described here. Known: UP1 = =6 V, U0 = 0.7 V, Rl = 6 kOhm (obtained by measurement, see Fig. 1, b). Using formula (5) we obtain K and - 4.78. Load R11 is three resistors connected in parallel R1. Transistor V6 maybe KT603A with a coefficient h21E, equal to 10 ; The input resistance of such an emitter follower will be about 20 kOhm.

To ignore the influence of the emitter follower input resistance V6, let's take a resistor R3 resistance 2.2 kOhm, then from formula (5) it follows that the resistor resistance R2 should be 460 ohms.

To ensure the rated voltage at the output of the stabilizer, taking into account the voltage drop at the emitter-base junction of the transistor V6 it is necessary that the resistor R3 the voltage drop was 6.7 V. To do this, you need to set the transistor collector current V5, equal to 3 mA, applying a bias voltage of 2.1 V to its base. Voltage drop across the diodes VI..UZ will be 2.1 V, so the resistance R1 - 0. Any silicon diodes can be used, but diodes are best.

KD503A, through which stable current drain field effect transistor V4. The most suitable transistor is KL302A with an initial drain current Ico = 10 mA. Stabilizer supply voltage Ua is chosen so large that all transistors operate in the active region. To do this, it is necessary to fulfill the condition

U n > kU Kn + ITo(R, + R 3),(6)

Where U Kn - transistor saturation voltage V5 for a given I k, To- safety factor (1.5…2.0).

For our example Ua should be more than 8.13 V. Let's choose 9 V. This concludes the calculation of the stabilizer.

The timing characteristics of pulsed devices of the type considered can be controlled by short-circuiting part of the current i 0 to the common wire. Current i 1 , charging capacitor C1, decreases by the value withdrawn from the point b current i 2. Then, using formula (2), transformed into the formula

Where i 1 - current charging the capacitor C1, we obtain a simplified expression for the dependence of the front delay time on the current connected to the common wire:

In the pulse edge delay device according to the diagram in Fig. 9 delay time is controlled by the voltage supplied to the modulating input. This voltage can be either constant (slowly changing) or pulsating.

A transistor serves as a current conductor VI, the current through which is determined by the control voltage and resistor values R1, R2. Resistor R1 plays the role of a base current limiter (transistor VI. Resistor R2 affects the linearity of the modulation characteristic and dynamic range control voltages.

Rice. 9. Circuit diagram of a pulse edge delay device with a delay time modulator

Current i 1 is limited by the requirement to ensure the operation of the transistor V8 in key mode. In practice this means that

i 1Max= i 0 - ibn. (9)

Here i 1 m ax - maximum value drained current, f c n - saturation current of the transistor base V8, equal

Rice. 10. Functional diagrams of pulse edge delay devices with in various ways delay time modulation: A- control voltage; b- control current

From formulas (9) and (10) the maximum value of the withdrawn current is determined:

(11)

For K217 series microcircuits i 1max = 0.8b mA. Based on the known value of the maximum withdrawn current, the down conductor can be calculated.

Modulation of the control voltage in the device according to the diagram in Fig. 10, A carried out with resistance R1=/=O And R2=/=0. In this case, the spread of transistor parameters practically does not affect the value of the withdrawn current. When choosing a transistor with a coefficient h 21E >10, when the base current can be neglected, the calculation of the modulator is simplified. In this case, the withdrawn current, approximately equal to the emitter current, is equal to

(12)

where U be is the base-emitter voltage of the transistor: for silicon transistors you can take: 0.7 V, for germanium transistors - 0.4 V.

Resistor resistance R2 can be calculated using formula (12).

When calculating a transistor-current collector of this modulation option, it should be borne in mind that with increasing resistor resistance R2 The drain transistor may be in saturation. This must be checked based on the condition (see Fig. 9)

(13)

Modulation by control current according to the diagram in Fig. 10, b carried out by the high resistance of resistor R1. In this case, the base current of the transistor V2 equals

i 6 =Ucontrol/R 1 , (14)

and the collector current V2, he is equal

i 1 = h 21Eib. (15)

From formulas (14) and (15) it follows that the withdrawn current depends on the control voltage:

To calculate the resistance of a resistor R1 it is necessary to substitute into formula (16): U ex. = U ulR. mAK s - maximum value of the control voltage, i 1 = i 1Max - maximum value of the withdrawn current from (11), h21E= = h 21emaks - maximum value h 2 i 3 current drain transistor.

But this modulation method has a significant drawback associated with current variability i 1 due to parameter variation h21E current drain transistor.

If temperature stabilization of down conductors is necessary, calculations are carried out similarly to calculations of temperature stabilization of amplifier stages.

When using the considered methods of modulating the timing characteristics of pulsed devices, it is possible to design:

voltage - PWM converter (pulse width modulation) from a standby multivibrator or from a pulse shaper of a given duration;

voltage - VIM converter (time-pulse modulation) from delay devices;

voltage-frequency converter. from a self-oscillating multivibrator, but using current leads in each arm of the multivibrator.

These converters produce signals with a spectrum, the width of which can be adjusted by voltage. Therefore, they can also find application in the design of electric musical instruments.

Described pulse devices can be designed on the logical elements of DTL series microcircuits: K217, K121, K194. From TTL microcircuits you can use the K133, K155, K158 and others series. The ones disassembled here differ favorably from previously published similar devices in that they contain fewer discrete components per logical element, and therefore their installation is reduced to a minimum.