Amplifier with current output. Operational amplifiers. Remote control switching circuits

The C-2820 is a fully balanced, quality, quality preamp with three sets of RCAs and three balanced inputs for a set of RCAs.

The input signal enters the input buffer channel after switching through the relay. Relay switching is pure mechanical switching, which can eliminate sound and paint.

The input buffer accepts the diamond current differential amplification module. This circuit has a wide frequency response, positive sound, no emotional color, and boiled water. Its advantage is that it acts as a matching impedance before and after the edge, so that the main amplifier circuit can operate without interference.

Circuit Features:

1. positive, negative half-cycle signal, whether AC or DC, the output amplitude is equal.

2. No crossover distortion, when the signal frequency changes from 2 Hz to 500 kHz, the oscilloscope does not notice any crossover. The more distortion, even if the output signal is very small, there is no crossover distortion, which because of the four pieces of tubse is Ube = 0.7 in silicon, and the Ube of each tube is built on a smart connection. Formed by mutual restraint, it is stable and reliable and is not influenced by the outside world and people.

3. The quiescent current can be determined by the voltage and two resistors connected to the power supply. The DC size does not need to be put into operation after turning on the power.

4. The output 0 potential is stable, when the tube pairing error is even 20% difference, when the +-supply voltage difference is 5V, the output can basically maintain 0V.

The main amplification circuit accepts full symmetrical current amplification.

The balanced output accepts four sets of identical analog modules. As a balanced positive and negative output, true full balance must use four channels. The circuit takes the form of current amplification rather than voltage amplification. The real amplified sound is really nice, the sound is clear and bright, and the high frequency is straight to the hairless cloud, the low frequency is powerful but flexible, the amplifier and 99% of the former use of current amplifier circuit.

The sound is ethereal and beautiful, elegant and noble, dynamic, the effect is 100% like the pre-level of 100000 golden scorpion, the sound effect is first-class. The new chassis design is beautiful and simple, and it is also equipped with a remote control that can remotely control the volume and six groups of signal inputs for more convenient use.

Machine Features:

1. advanced products, customization level plays an important role in sound style.

C-2820 preamplifier, sound characteristics: dynamic, wide sound field, mid-foot, rich in detail, high-frequency, bright and penetrating, with a little extravagance, bass is fierce, deep immersion, sound calm atmosphere, voice merciful and natural, and the energy is full of energy.

3, power transformer:

The high quality imported transformer is used, sealed with epoxy resin, and the magnetic leakage is shielded by a beautiful cow case, which can improve the sound clarity. A transformer without shielding, a case with cows, has an effect on the machine, which is not good, especially a pre-stage one. It feeds from Shuangniu and feeds from 2 cattle. Now many fans really like it. If you don't care about the cost, the effect is definitely better than 2, the sound effect brought by the double cow is obvious, the sound is full, and transparent.

4220 V Power consumption

The power source is very important. The current power supply is very dirty. Every home has an air conditioner, refrigerator, and color TV, which will pollute the city's electricity and make the sound unclean. For this reason, we have added "DC Green" in the front stage. “The design, exclusive research and development, the effect is first-class.

5, supply voltage regulator

A two-transformer dual regulation system is used to separate and feed most power supplies into a balanced circuit. The gain line part is further regulated by electronic filtering technology, which can make the power supply ripple several orders of magnitude lower. After professional design, it will have professional effects.

6: Input switching can be remote control, volume can be remote control

Three sets of RCA inputs, three sets of balanced inputs, can be switched remotely at will, or the front panel can be manually turned on. The volume can be controlled remotely while sitting on the sofa. Everything is people-oriented, tactful and practical.

7: Adopting solid aluminum chassis

Panel 10 mm. The top and bottom covers are 4 mm, which is almost twice their thickness. It feels heavy in the hand, plus the 4 solid aluminum CNC machining shock absorber legs. Then use the O-ring as a buffer to make the upper chassis shock absorber effect and sound to a higher level.

Key Features:

Frequency characteristics: 20Hz~20kHz (+0-0.2dB)

3Hz ~ 200kHz (+0-3dB)

Full Distortion Range: 0.004%

Input sensitivity: 250mV/40K (balanced) 250mV/20K (RCA)

Nominal input level: 2V (audio signal)

Nominal Output Level: 2V/50Ω,(Beep)

Maximum output level: 10V (beep)

Signal-to-noise ratio: 115db

Minimum load resistance: 600 ohms

Power supply: AC-220V 50/60Hz

Dimensions: 430mm (width) X310mm (deep) X110mm (height) (excluding foot) X310mm

Weight: 18 kg,

From a package of 22 kg

As follows from (2.12), the current output, i.e. high output resistance, implemented with negative current feedback (Fig. 3.3, A) or positive voltage feedback (Fig. 3.3, b). Let's find the expression for the output (relative to the load) resistance of the amplifier with an ungrounded load (Fig. 3.3, A), using relation (2.12), where in relation to the circuit in Fig. 3.3,

From this it can be seen that the output resistance in the circuit of Fig. 3.3, A under the action of negative feedback, the current turns out to be several times greater than the same resistance measured in the absence (i.e., open) of the OS. Input resistances in the circuit Fig. 3.3, A in relation to signal sources and are the same as in the diagrams in Fig. 3.1, A And b. Taking into account relation (3.2), we find the expression for the load voltage for inverting (), non-inverting () and differential amplifiers:

As can be seen from these expressions, the voltage across the load is directly proportional to the load resistance, and the current in the load does not depend (within the accepted assumptions) on, which is a sign of the current output. Subtraction error in the circuit in Fig. 3.3, A, in contrast to the diagram in Fig. 3.1, V, does not depend on the accuracy of the resistances of external resistors, but is determined only by the differential properties of the op-amp itself (common-mode signal attenuation coefficient).

To simplify the analysis of the amplifier circuit with a grounded load (Fig. 3.3, b), we transform it into a one-loop circuit, as shown in Fig. 3.3, V, where the operational amplifier with a negative feedback loop is represented by an equivalent amplifier (EA) with finite gains

Since in this circuit there is only one feedback loop, and it is positive and voltage (on the load side), to determine the output conductivity we use expression (2.12), where, a. If the condition is met, then the output conductivity becomes zero (with an ideal op-amp):

Considering that the direct transmission coefficients (with the OS loop open) from the output are respectively equal

Submitting your good work to the knowledge base is easy. Use the form below

Students, graduate students, young scientists who use the knowledge base in their studies and work will be very grateful to you.

Http://www.allbest.ru/

Amplifiers with current output

Considering that the direct transmission coefficients (with the OS loop open) from the output are respectively equal

from (2.10) we obtain expressions for the transmission coefficients of an amplifier with a grounded load

Load voltage expressions for inverting (), non-inverting () and differential amplifiers

confirm that shown in Fig. 3.3, b The circuit is a current output amplifier circuit.

Phase shifters

The phase shifter allows you to set the required phase shift at a certain frequency without changing the module of the transfer function. Knowing the transfer functions from the inverting and non-inverting inputs of the op-amp, covered by negative feedback, we will find expressions for the transfer functions of the phase shifters, the diagrams of which are shown in Fig. 3.4, A And b:

If the condition is met, then these functions will take the form

according to which we obtain the following expressions for the amplitude-frequency and phase-frequency characteristics:

phase shifter current amplifier grounded

Thus, the diagrams in Fig. 3.4 can be used as phase correctors, in which the modulus of the transfer function in a wide frequency range (where the operational amplifier can be considered ideal) does not depend on frequency.

Integrators

Since the integration operation is implemented with a linear charge and discharge of a capacitor (which requires a current source with a sufficiently large, ideally infinite, resistance), the integrator circuit is obtained from an amplifier circuit with a current output (see Fig. 3.3, A And b), if instead you turn on the capacitor, as shown in Fig. 3.5. To have an unbalanced low-impedance output, the integrator output signal is taken not from the capacitor, but from the output of the op-amp, but the voltage at the output of the op-amp and the voltage at the capacitor (up to a scale factor) coincide only in the case when the input signal in the circuit of Fig. 3.5, A is supplied to the inverting input of the op-amp, and in the circuit of Fig. 3.5, b- to the non-inverting input.

Since in the diagram Fig. 3.5, A(the voltage in node 1 is close to zero), and in the circuit of Fig. 3.5, b(i.e. amplified by a non-inverting amplifier), expressions for the output voltage of the integrators can be obtained from (3.5) and (3.6) when replaced by:

Assuming that in these expressions p- Laplace operator, let's move from images to originals:

Having expressions for the transfer functions of the inverting and non-inverting integrators

Let's plot (Fig. 3.6) their amplitude-frequency and phase-frequency characteristics

where in the diagram Fig. 3.5, A; in the diagram of Fig. 3.5, b. The time constant is set based on the frequency range of the input signal and the requirements for the output voltage.

The above relations were obtained under the assumption that the operational amplifier is ideal, and in the non-inverting integrator circuit also under the condition. If in the diagram Fig. 3.5, A take into account the finiteness of the op-amp gain, and in the circuit of Fig. 3.5, b- possible deviations of resistances from their calculated values (,), then the transfer functions of the integrators will take the form

where is the integration error in the circuit shown in Fig. 3.5, A significantly less error in the diagram in Fig. 3.5, b. Deviations of frequency characteristics from ideal ones caused by an error are shown by the dotted line in Fig. 3.6. When constructing differential or non-inverting integrators with a small error, the circuit in Fig. is used. 3.5, A with the addition of a differential one at its input (see Fig. 3.1, V) or inverting (see Fig. 3.1, A), and in the general case summing (see Fig. 3.2, b) amplifier.

The frequency properties of the op-amp in the high-frequency region affect the integration accuracy in the case of a small integrator time constant. Integration errors due to the finite resistance and non-zero op-amp are negligible if the external resistors and capacitor values are properly selected. The main problem when constructing integrators is the drift of the op-amp zero. Actually shown in Fig. 3.5 circuits can only be used when they, as part of a more complex circuit, are covered by negative DC feedback. If such conditions are absent, then integrators with reset are used (see Section 8.4).

Differentiators

The inverting differentiator circuit is obtained from the corresponding integrator circuit (see Fig. 3.5, A) when rearranging the resistor and capacitor, as shown in Fig. 3.7, A.

Like the integrator, the differentiator is described by the transfer function and frequency characteristics

written here for the case of an ideal op-amp. The finiteness of the gain of the operational amplifier and its frequency properties affect the differentiator in the high frequency region (shown by the dotted line in Fig. 3.8). However, the main differentiation error arises from high-frequency electrical noise of the operational amplifier, since in the region of sufficiently high frequencies negative feedback has practically no effect (low capacitor resistance) and the noise voltage at the op-amp output is significant. Therefore, in reality the diagram in Fig. 3.7, A can only work as part of a more complex circuit that has a fairly deep overall negative feedback in the high frequency region.

In order to reduce the noise output voltage, a resistor is connected in series with the capacitor (Fig. 3.7, b), which increases the depth of negative feedback at high frequencies. In this case, the expressions for the transfer function and frequency characteristics take the following form:

where the differentiation error depends on frequency. By rationally selecting the resistance value, it can be made acceptable in the operating frequency range, while at the same time ensuring a sufficiently low level of high-frequency noise output voltage. Deviation of frequency characteristics of a real differentiator (Fig. 3.7, b) from the characteristics of the ideal (Fig. 3.7, A) shown in Fig. 3.8 dotted line. Type of distortion of characteristics in the circuit Fig. 3.7, b the same as with a non-ideal op-amp in the circuit of Fig. 3.7, A, but the range of operating frequencies in the circuit of a real integrator is much smaller (at the same time, we recall, the level of high-frequency noise is also lower).

A non-inverting, differential or multi-input differentiator can be built based on one of the considered circuits by connecting an inverter, differential amplifier or adder to its input.

Posted on Allbest.ru

...

5. DC AMPLIFIERS

5.1. General information

Direct current amplifiers (DCA) are devices designed to amplify slowly varying signals down to zero frequency. Figure 5.1 shows the frequency response of the UPT.

Figure 5.1. Frequency response UPT

To transmit signals of frequencies close to zero, the UPT uses direct (galvanic) coupling between cascades. However, such a connection leads to the need to solve specific problems:

◆ coordination of potential levels in neighboring cascades;

◆ reducing drift (instability) of the output voltage or current level.

5.2. Methods for constructing a UPT

The main problem faced by UPT developers is zero drift. Zero (zero level) drift is a spontaneous deviation of the voltage or current at the output of the UPT from the initial value. Since zero drift is observed even in the absence of a signal at the input of the UPT, it cannot be distinguished from the true signal.

The physical reasons causing zero drift in the UPT include:

◆ instability of power supplies;

◆ temporary instability (“aging”) of transistor and resistor parameters;

◆ temperature instability of transistor and resistor parameters;

◆ low-frequency noise;

◆ interference and interference.

The greatest instability is caused by the temperature factor. The situation is aggravated by the presence of galvanic coupling between the cascades, which transmits slow signal changes well, which leads to the effect of cascading temperature instabilities of the cascades from input to output.

Since temperature changes in the parameters of amplifying elements are of a natural nature (see subsections 2.2 and 2.10), they can be compensated to some extent by the same methods as in amplifiers of harmonic signals.

Absolute zero drift Δ U out is called the maximum spontaneous deviation of the output voltage of the UPT with a closed input over a certain period of time. The quality of the UPT is assessed by the zero drift voltage referred to the amplifier input:

e dr = Δ U out/K U.

The zero drift applied to the input is equivalent to a false input signal; it limits the minimum input signal, i.e. determines the sensitivity of the UPT.

In order to reduce zero drift, the UPT uses:

◆ deep environmental protection;

◆ thermal compensating elements;

◆ conversion of direct current into alternating current, its amplification and subsequent detection;

◆ construction of the UPT according to a balanced scheme.

Direct amplification UPT, in fact, they are ordinary multi-stage amplifiers with direct coupling. An amplifier, the circuit of which is shown in Figure 3.4, can be used as a UPT.

In this amplifier, resistors R e1, R e2 and R e3, in addition to creating local and general OOS circuits, provide the necessary bias voltage in their stages. In a multistage UPT, it is possible to ensure the required DC mode of transistors by sequentially increasing the emitter potentials from input to output, which is due to the direct interstage collector-emitter connection; the collector potentials also increase from input to output. It is possible to ensure the UPT cascade mode by reducing R to from input to output, however, in both cases the consequence will be a decrease in the gain of the UPT.

In multi-stage direct amplification amplifiers, partial compensation of zero drift can occur. Thus, a positive increase in the collector current of the first transistor will cause a negative increase in the base current and, consequently, the collector current of the second transistor. In practice, complete compensation of zero drift is not achievable even for one temperature point; however, in a UPT with an even number of cascades, its reduction is observed.

Due to the fact that this UPT has a unipolar power supply, there is some constant potential at its input and output, which does not allow connecting a low-impedance signal source and load directly between them and the common wire. In this case, a bridge circuit is used with the inclusion of R G and R n in the diagonal of the input and output bridges (Figure 5.2).

Figure 5.2. Bridge circuit for connecting the signal source and load in the UPT

To calculate the frequency and time characteristics of the UPT with direct amplification, you can use the materials of subsections 2.5 and 3.3, as well as subsection 2.9 in the case of constructing the UPT on the PT.

For the purpose of matching potentials, transistors of different conductivities are used; for better temperature compensation, diodes and zener diodes are used. The use of a bipolar power supply allows you to directly connect the signal source and load to the UPT, because in this case, zero potentials are provided at its input and output. These measures are implemented in the UPT scheme shown in Figure 5.3.

Figure 5.3. Two-stage UPT

Direct amplification UPTs have a large temperature drift ( e dr amounts to units of millivolts per degree). In addition to temperature drift in such UPTs, time drift, instability of power supplies and low-frequency noise have a significant impact.

The noted shortcomings have been largely overcome V UPT with signal conversion (modulation). Figure 5.4 shows a block diagram of the UPT with the conversion of direct current to alternating current and provides voltage diagrams that explain the principle of its operation.

DC input signal U in converted into an AC voltage signal proportional to it using a modulator M, then amplified by a conventional harmonic amplifier U, and then the demodulator DM converted to DC voltage signal U n. Since in AC amplifiers the zero drift is not transmitted from stage to stage (due to the presence of separation capacitors between the stages), then in this UPT the minimum zero drift is realized.

Figure 5.4. Block diagram of the UPT with signal conversion

As a modulator, you can use controlled key circuits, usually made on a PT. The simplest demodulator is a conventional full-wave rectifier with an output filter. It should be noted that there is a wide variety of circuit designs for both modulators and demodulators, the consideration of which is not allowed by the limited scope of this manual.

The disadvantages of UPT with signal conversion include the problem of implementing low-level input signal modulators and the increased complexity of the circuit.

It is possible to achieve a significant improvement in the electrical, operational and weight-size parameters of UPTs by constructing them on the basis of balanced circuits.

5.3. Differential Amplifiers

Currently, the most widely used UPTs are based on differential (parallel-balanced or difference) cascades. Such amplifiers are simply sold in the form of monolithic ICs and are widely produced by industry (KT118UD, KR198UT1, etc.). Figure 5.5 shows a schematic diagram of the simplest version of a differential amplifier (DA) on a BT.

Figure 5.5. Remote control diagram

Any remote control is performed on the principle of a balanced bridge, two arms of which are formed by resistors R k1 and R k2, and the other two by transistors VT 1 and VT 2. The load resistance Rn is included in the diagonal of the bridge. The POOST circuit resistors R OS1 and R OS2 are usually small or absent altogether, so we can assume that the resistor R e is connected to the emitters of the transistors.

Bipolar power supply allows you to do without bridge circuits at the inputs (outputs) of the remote control by reducing the potentials of the bases (collectors) to the potential of the common bus.

Let's consider the operation of the remote control for the main operating mode - differential. Through action U in 1 transistor VT 1 opens slightly, and its emitter current receives an increment Δ I uh 1, and due to the action U in 2 transistor VT 2 closes, and its emitter current receives a negative increment –Δ I uh 2. Consequently, the resulting increase in current in the resistor circuit R e with perfectly symmetrical arms is close to zero and, therefore, there is no negative feedback for the differential signal.

When analyzing the remote control, two arms are identified, which are cascades with OE, in the common circuit of the emitters of the transistors a common resistor R e is included, which sets their total current. In this regard, it seems possible to use the relationships of subsections 2.5 and 2.12 when calculating the frequency and time characteristics of the remote control, taking into account the comments given in subsection 4.4. For example, the differential signal gain K U differential will be equal in case of symmetry of the shoulders (see subsection 4.4) K U differential=2· K U pl=K 0, i.e. the differential gain is equal to the gain of the cascade with the OE.

The remote control is distinguished by a small zero drift and a high gain of the differential (anti-phase) signal K U differential and a high common-mode noise suppression coefficient, i.e. low common mode transmission ratio K U sf.

To ensure the quality performance of these functions, two basic requirements must be met. The first of them is to ensure the symmetry of both arms of the remote control. Microelectronics has made it possible to get closer to fulfilling this requirement, since only in a monolithic IC, closely spaced elements actually have almost the same parameters with the same response to the effects of temperature, aging, etc.

The second requirement is to provide deep feedback for the common mode signal. The common-mode signal for the remote control is interference and interference arriving at the inputs in phase. Since R e creates a deep POOST for both arms of the remote control, then for a common-mode signal there will be a significant decrease in the transmission coefficients of the cascades with OE forming these arms.

The gain of each arm for a common mode signal can be represented as K 0OS cascade with OE with deep OOS. According to subsection 3.2 we have:

K U sf 1 ≈ R to 1 /Re,

K U sf 2 ≈ R to 2 /Re.

Now we can write for K U sf total remote control:

K U sf ≈ Δ R to/Re,

where Δ R to= |R to 1 – R to 2 |.

To evaluate the suppression of a common-mode signal, a common-mode signal attenuation coefficient (CMRR) is introduced, equal to the ratio of the modules of the transmission coefficients of the differential and common-mode signals.

From the above it follows that increasing the CMRR is possible by reducing the spread of resistor values in the collector circuits (in monolithic ICs - no more than 3%) and by increasing R e. However, the increase R e requires an increase in the power supply voltage (which will inevitably lead to an increase in the dissipated thermal power in the remote control), and is not always possible due to technological difficulties in implementing large-value resistors in monolithic ICs.

This problem can be solved by using the electronic equivalent of a large value resistor, which is a stable current source (SCS), the circuit options of which are shown in Figure 5.6.

Figure 5.6. IST on BT and PT

The IST is connected instead of R e (see Figure 5.5), and the specified current and thermal stability are provided by the elements R 1, R 2, R e and VD 1 (Figure 5.6a), and R 1 (Figure 5.6b). For real conditions, the IST is the equivalent of a resistance for a changing signal with a nominal value of up to several megohms, and in rest mode - on the order of several kilo-ohms, which makes the remote control economical in terms of power.

The use of IST makes it possible to implement the remote control in the form of an economical IC, with a CMRR of about 100 dB.

When using a PT, the nature of the design of the remote control does not change; you only need to take into account the features of power supply and thermal stabilization of the PT.

5.4. Remote control switching circuits

There are four remote control switching circuits: symmetrical input and output, unbalanced input and symmetrical output, symmetrical input and unbalanced output, unbalanced input and output.

Remote control circuit diagram balanced input and output is shown in Figure 5.7 and does not require any special comments; this connection circuit is used when cascading remote control.

Figure 5.7. Switching diagram of the remote control “symmetrical input and output”

Remote control circuit diagram unbalanced input and balanced output discussed earlier (see Figure 4.9).

Remote control circuit diagram balanced input and unbalanced output is shown in Figure 5.8.

Figure 5.8. Switching diagram of the remote control “balanced input - unbalanced output”

This remote control switching circuit is used if it is necessary to transition from a symmetrical signal source (or a symmetrical transmission path) to an asymmetrical load (an asymmetrical transmission path). It is easy to show that the differential gain with such a connection will be equal to half K U differential with symmetrical load. Instead of resistors Rk, transistors are often used in the remote control, performing the functions of dynamic loads. In the considered option for switching on the remote control, it is advisable to use the so-called current mirror , formed by transistors VT 3 and VT 4 (Figure 5.9).

Figure 5.9. Remote control circuit with current mirror

When a positive half-wave of a harmonic signal is applied to the base of transistor VT 1 U in 1, in the circuit of transistor VT 3 (connected according to the diode circuit) a current increment Δ occurs I to 1. Due to this current, a voltage increment occurs between the base and emitter VT 3, which is an increment in the input voltage for the transistor VT 4. Thus, in the collector-emitter circuit VT 4 there is a current increase almost equal to Δ I to 1, since in the remote control the shoulders are symmetrical. At the moment in time under consideration, a negative half-wave of the input harmonic signal is supplied to the base of transistor VT 2 U in 2. Consequently, a negative current increment Δ appeared in the circuit of its collector I to 2. In this case, the increment of the remote control load current is equal to Δ I to 1 +Δ I to 2, i.e. A remote control with a current reflector provides greater differential signal amplification. It should also be noted that for the considered remote control option in quiescent mode, the load current is zero.

At unbalanced input and output The operation of the remote control is, in principle, no different from the case of unbalanced input - symmetrical output. Depending on which arm the output signal is taken from, it is possible to obtain an in-phase or anti-phase output signal, as is obtained in a phase-inverted stage based on a remote control (see subsection 4.4).

5.5. Remote control accuracy parameters

Voltage U cm is generated mainly by the spread of the reverse currents of the emitter junctions I ebo 1 and I ebo 2 (U" cm), and the spread of resistor values R k1 and R k2 ( U" cm). For these voltages we can write:

U" cm = φ T ln( I ebo 1 /I ebo 2),

U" cm= 2· φ T·Δ R to/R to.

Addiction U cm on temperature appears to be another precision parameter - temperature sensitivity. Temperature sensitivity dU cm/dT has a dimension of μV/deg and is defined as the difference between the TKN of the emitter junctions of the transistors of the arms and decreases in proportion to the decrease U cm.

The next precision parameter of the remote control is the bias current Δ I input, which is the imbalance (difference) of input currents (transistor base currents). Flowing through the signal source resistance Rg, the bias current creates a voltage drop across it, the effect of which is equivalent to a false differential signal. The bias current can be represented as

Δ I input = I e01/ H 21E1 – I e02/ H 21E2.

Average input current I input wed is also an accuracy parameter of the remote control. It can be represented as

I input wed = (I b01 + I b02)/2 = I e0 /2 H 21E.

Flowing through R g, current I input wed creates a voltage drop across it, acting as a common-mode input signal. Although weakened in K Usf times, it will still cause a potential imbalance at the output of the remote control.

The temperature dependences of the bias current and the average input current can be taken into account through the temperature dependence H 21E. Note that usually I input wed>Δ I input.

In the DC remote control, the main accuracy parameter is U cm, which is usually greater than in the remote control on BT.

Currently, remote controls represent the main basic stage of analog ICs; in particular, the remote control is the input stage of any operational amplifier.

The problem of a high-quality, but simple and cheap headphone amplifier remains relevant. Personally, I have a problem with the Bayerdynamic 880 250 Ohm headphones. They work cleanly, but soullessly, like a monitor, you don’t want to listen to them. Therefore, I decided to assemble and test the amplifier described in Radiohobby magazine No. 1 for 2011. Read below to see what came of it.

The assembled device looks like this:

To begin with, everything written below is my subjective opinion, yours may be the opposite. This amplifier is unusual in that it has a very high output impedance; by the way, its author is not Lipavsky, but Safronov. I wrote personally to the author regarding issues of authorship and the operation of the amplifier, but, alas, I did not receive any answer...

I made the layout of the printed circuit board so that, if necessary, you can place transistors in pairs on small heat sinks, and added space for “sandwiches”. I think that it is necessary to install sockets under the microcircuits in order to try out different options, and even if the microcircuit is accidentally damaged, replacement will not cause problems. As usual, I try to make the paths as thick as possible.
On the board, jumpers are shown with thick red lines, purple holes for technology. purposes - combination for transfer of part designations using the LUT method.
I forgot to draw the power capacitance on the diagram (the circuit board and photo are shown), I put 220 uF 16 V each. That’s what I wanted and there was a place.

And the power supply diagram.

Fragment excluded. Our magazine exists on donations from readers. The full version of this article is available only

I made some minor minor amateur radio changes, such as the output voltage of the mains transformer being excessive, but I didn't have anything else at hand.
The power supply must be stabilized - noise and ripple will be picked up by the amplifier.
Transistors and stabilizers get quite hot. I had to install small heat sinks in the power supply. I installed transistors KT814 and KT815 in the amplifier. Since their heating with a power supply of ±9 V and a current of 60 mA is approximately 60...70 degrees, I refused to use heat sinks for them. In general, the setup is very simple - you just need to set the desired quiescent current and you can (optionally) minimize the constant voltage at the output. By the way, both Safronov and Lipavsky made the same gross mistake in the circuit - the wrong value of resistor R3. As they say, small lies breed great mistrust...
The quiescent current is set by resistor R3, more resistance means less current. Set the variable to 220 kOhm and, reducing its resistance, control the voltage on R6 and R7; for 60 mA at 51 Ohm it should be 3 V.
The original sources recommend achieving zero voltage at the output by selecting resistors R2 and R4. I declare that this is impossible to do; balance can be achieved by soldering (after warming up the amplifier) shunt resistances in parallel with resistors R6 or R7. In addition, we had to add a resistor in parallel with the headphones to reduce the DC component, otherwise it is impossible to achieve zero at the output without headphones. After switching on, the constant voltage at the output begins to float back and forth over a wide range, so I recommend connecting headphones after the end of transient processes - after about half a minute. Pay attention to flushing the board after soldering!
I tested with a quiescent current of 60 mA, although 30 mA would be enough for high-impedance headphones. I assume that the amplifier must deliver at least 100 mW to the load, so I set the power supply to plus or minus 9 Volts.

Impressions from listening

Source - old Marantz CD player, various recordings, mostly unlicensed, genre mainly instrumental and orchestra. Headphones - Bayerdinnamik 880, Grado 125 and a little Koss 2000.

The amplifier works no worse than the built-in CD player, and with headphones with an impedance of 250 Ohms it can work much louder (for 32 Ohms the volume of a CD player is quite enough). The noises are very small and in a living room are only audible in the absence of a phonogram, and even then, if you strain your ears, during operation the amplifier’s own noises are not audible at all. The bass performs surprisingly well (my hypothesis for why is below). But at high frequencies there is some harshness, excessive brightness, turning into a metallic sound. I tried a number of CHEAP OUs, I can’t vouch for their “pedigree”. These are: 4558, 4556, LM358, TL082, 5532. Since I changed them several times without seeing the markings, this can be considered a “blind examination”. The difference between them is very small, but it seemed to me that 5532 worked a little better.
It must be said that the shortcomings of phonograms and recordings are very noticeable, which can be attributed to the advantages of the tract, and blanks burned from apecu sound the worst of all, no matter what they try to talk about “lossless copying”, this is not news to me for a long time.
After reflection, I came to the conclusion that the “sounding features of a current amplifier” may be due to the fact that the voltage across the load is directly proportional to its resistance (at a stable current due to the high output resistance), which means that an increase in impedance leads to an increase in voltage and volume . I explain the noticeable increase in bass by an increase in Z and, accordingly, voltage at low frequencies. It should be noted that with Bayerdynamics the increase in Z at the resonant frequency is very small and, unlike LF speakers, it is a percentage, not times. I don’t know how it will be with other headphones. There is no muttering, not at all. Another thing is that with different headphones, the sound of the amplifier will be different, for example, KOSS 2000 sounded bad. Nevertheless, this amplifier is friendly with Bayer speakers. I didn’t make a palace (box) for the amplifier; I want to make a tube amplifier and compare.
In general, the amplifier works well and is quite suitable even for budget block equipment; it is worth noting the cheap cost of its parts and ease of setup. My recommendations are that for low-impedance headphones you can slightly reduce the supply voltage and try to increase the current; for high-impedance headphones - increase the supply voltage and reduce the current, I made a universal option. Due to the excess of highs, I tried to adjust the R5C5 circuit, this made the sound somewhat softer, the sibilance became non-irritating. Probably, you need to install a simple low-pass filter in the form of an RC chain at the input.
The most surprising thing is that with this amplifier, Bayerdynamics began to outplay our family Grado headphones, after all, I was right - it was necessary to do recabling...

Listening Impressions 2

The source is a computer with an ASUS Xonar sound card, which I modified a little - I installed a “cool” and no longer cheap AD8066 chip as the output. It immediately gave a noticeable increase in quality compared to the standard microcircuit. Recordings in “lossless” formats and mp3 320kbps. Headphones - Bayerdynamics only. The software players are different, all sorts of effects and equalizers are, of course, turned off.
... To be honest, I don’t want to write, as they say, another class. The headphones and amplifier have nothing to do with it, the source is to blame - the computer. Compared to CD the sound is simply poor. The pleasantness of the lows has disappeared, the stage is compressed, the air has disappeared, etc. Of course, not everything is so bad, it just depends on what you compare it to. If you only listen to the computer and other gadgets, you may well like it; the high frequencies, by the way, will be even softer here. The sensitivity of the amplifier is just enough with a small margin. However, every cloud has a silver lining; for a computer this amplifier is no longer a weak link, but for some notebook it is already more than enough.