Vasilich amplifier with N-channel output stage by Alexey Nikitin. Circuit design of output stages of power amplifiers Current mirror in the output stage of the UMZCH
Recently, more and more often, many companies and radio amateurs are using powerful field-effect transistors with an induced channel and an insulated gate in their designs. However, it is still not easy to purchase complementary pairs of field-effect transistors of sufficient power, so radio amateurs are looking for UMZCH circuits that use powerful transistors with channels of the same conductivity. The magazine “Radio” published several such designs. The author proposes another one, but with a structure slightly different from a number of circuits common in UMZCH designs.
Technical Parameters:
Rated output power into 8 ohm load: 24 W
Rated output power into 16 Ohm load: 18 W
Harmonic distortion at rated power into 8 ohm load: 0.05%
Harmonic distortion at rated power into 16 ohm load: 0.03%
Sensitivity: 0.7V
Gain: 26 dB
For the last three decades, the classic transistor UMZCH has used a differential cascade. It is necessary to compare the input signal with the output signal returning through the OOS circuit, as well as to stabilize the “zero” at the amplifier output (in most cases, the power supply is bipolar, and the load is connected directly, without an isolating capacitor). The second is the voltage amplification stage - a driver that provides the full amplitude of the voltage required for the subsequent current amplifier on bipolar transistors. Since this cascade is relatively low-current, the current amplifier (voltage follower) consists of two or three pairs of composite complementary transistors. As a result, after the differential stage, the signal passes through another three, four, or even five amplification stages with corresponding distortion in each of them and a delay. This is one of the reasons for the occurrence of dynamic distortions.
In the case of using powerful field-effect transistors, there is no need for multi-stage current amplification. However, to quickly recharge the gate-channel interelectrode capacitance of a field-effect transistor, a significant current is also required. To amplify audio signals, this current is usually much less, but in switching mode at high audio frequencies it turns out to be noticeable and amounts to tens of milliamps.
The UMZCH described below implements the concept of minimizing the number of cascades. At the amplifier input there is a cascade version of a differential stage on transistors VT2, VT3 and VT4, VT5, the load for which is applied to an active current source with a current mirror on transistors VT6, VT7. The current generator on VT1 sets the mode of the differential stage for direct current. The use of sequential connection of transistors in a cascade allows the use of transistors with a very high base current transfer coefficient, which are characterized by a small maximum voltage value (usually UKEmax = 15 V).
Between the negative power supply circuit of the amplifier (source VT14) and the bases of transistors VT4 and VT5, two zener diodes are connected, the role of which is played by the reverse-connected base-emitter transitions of transistors VT8, VT9. The sum of their stabilization voltages is slightly less than the maximum permissible gate-source voltage VT14, and this ensures protection of the powerful transistor.
In the output stage, the drain of the field-effect transistor VT14 is connected to the load through the switching diode VD5. Half-cycles of the minus-polarity signal are supplied through the diode to the load; half-cycles of the positive polarity do not pass through it, but are supplied through the transistor VT11 to control the gate of the field-effect transistor VT13, which opens only during these half-cycles.
Similar output stage circuits with a switching diode are known in the circuit design of bipolar transistor amplifiers as a stage with a dynamic load. These amplifiers operated in class B mode, i.e. without through quiescent current. In the described amplifier with field-effect transistors, there is also a transistor VT11, which performs several functions at once: a signal is received through it to control the gate VT13, and local feedback on the quiescent current is formed, stabilizing it. In addition, the thermal contact of transistors VT11 and VT13 stabilizes the temperature regime of the entire output stage. As a result, the output stage transistors operate in class AB mode, i.e. with a level of nonlinear distortion corresponding to most versions of push-pull stages. A voltage proportional to the quiescent current is removed from resistor R14 and diode VD5 and supplied to the base VT11. The VT10 transistor contains an active source of stable current, which is necessary for the operation of the output stage. It is a dynamic load for VT14 when it is active during the corresponding half-cycles of the signal. The composite zener diode formed by VD6 and VD7 limits the gate-source voltage of VT13, protecting the transistor from breakdown.
Such a two-channel UMZCH was assembled in the housing of the ROTEL RX-820 receiver to replace the UMZCH existing there. The plate heat sink is reinforced with metal steel struts to increase the effective area to 500 cm 2 . The oxide capacitors in the power supply were replaced with new ones with a total capacity of 12000 μF for a voltage of 35 V. Differential stages with active current sources (VT1-VT3) from the previous UMZCH were also used. The breadboards contain cascode continuations of the differential stage with current mirrors for each channel (VT4-VT9, R5 and R6) and active current sources for the output stages (VT10 of both channels) on a common board with common elements R9, VD3 and VD4. The VT10 transistors are pressed to the metal chassis with their back sides to avoid the need for insulating spacers. The output field-effect transistors are fixed to a common heat sink with an area of at least 500 cm2 through heat-conducting insulating pads with screws. Transistors VT11 of each channel are mounted directly on the terminals of transistors VT13 so as to ensure reliable thermal contact. The remaining parts of the output stages are mounted on the terminals of powerful transistors and mounting racks. Capacitors C5 and C6 are located in close proximity to the output transistors.
About the parts used. Transistors VT8 and VT9 can be replaced with zener diodes for a voltage of 7-8 V, operable at a low current (1 mA), transistors VT1-VT5 can be replaced with any of the KT502 or KT3107A, KT3107B, KT3107I series, and it is advisable to select them close in current transfer coefficient bases in pairs, VT6 and VT7 can be replaced with KT342 or KT3102 with letter indices A, B, in place of VT11 there can be any of the KT503 series. It is not worth replacing the D814A zener diodes (VD6 and VD7) with others, since the dynamic load current is approximately 20 mA, and the maximum current through the D814A zener diodes is 35 mA, so they are quite suitable. The inductor winding L1 is wound on resistor R16 and contains 15-20 turns of PEL 1.2 wire.
The establishment of each channel of the UMZCH begins with the drain outlet VT13 temporarily disconnected from the power circuit. Measure the emitter current of VT10 - it should be approximately 20 mA. Next, connect the drain of transistor VT13 to the power source through an ammeter to measure the quiescent current. It should not significantly exceed 120 mA, this indicates correct assembly and the serviceability of the parts. The quiescent current is regulated by selecting resistor R10. After turning it on, it should be immediately set to about 120 mA; after warming up for 20-30 minutes, it will decrease to 80-90 mA.
Possible self-excitation is eliminated by selecting capacitor C8 with a capacity of up to 5-10 pF. In the author's version, self-excitation arose due to a defective transistor VT13 in one of the channels. For other supply voltages, the heat sink area should be recalculated based on changes in the maximum power in one direction or another and ensure that the permissible parameters for the semiconductor devices used are not exceeded.
"Radio" No. 12, 2008
The amplifier proposed by the author is distinguished by the use of combined feedback (current and voltage across the load), which makes it possible to select the output resistance for a specific speaker in a wide range - from zero to tens of ohms. High linearity over the entire audio frequency band allows the use of such an UMZCH for broadband amplification of audio signals at a power of more than 100 W. The described amplifier has fairly high-quality parameters that contribute to good sound, and can be recommended for building high-quality sound reproduction systems. The ability to adjust the output impedance of the amplifier in the range from zero to several tens of ohms allows you to improve the sound quality of the speaker system. This makes the UMZCH ideal for working with a subwoofer made in a closed housing (without a bass reflex). Increasing the output impedance allows you to raise the level of low frequencies and reduce the lower cutoff frequency of the subwoofer. Sometimes the increased output impedance of the UMZCH contributes to the perception of the sound of the UMZCH-AS system, which is close to “soft tube sound”.
Maximum output power, W,
at a load of 4 Ohms 150
at a load of 8 Ohms 120
Harmonic distortion coefficient at an output power of 60 W at a frequency of 1 kHz, %,
no more than 0.005
Intermodulation distortion coefficient at frequencies 60 Hz/7 kHz, %, no more than 0.005
Intermodulation distortion coefficient at frequencies 18/19 kHz, %, no more than 0.005
Output voltage slew rate, V/µs, not less than 15
Output resistance, Ohm 0...20
Intermodulation distortion coefficient was measured using two methods: using the SMPTE method at frequencies of 60 Hz and 7 kHz with an amplitude ratio of 4:1, and also at frequencies of 18 and 19 kHz with an amplitude ratio of 1:1. The circuit diagram of the amplifier is shown in Fig. 1.
It is built according to a structure close to the structure of UMZCH Lin. The input differential stage on transistors VT3 and VT4 is loaded onto a current mirror on transistors VT1 and VT2 to obtain maximum gain, symmetry and rate of rise of the output voltage. Resistors R5 and R6 in the emitters increase the linearity of the cascade and its overload capacity, and also reduce the influence of the spread of transistor parameters. The current source on transistors VT5, VT6 (compared to a resistor, which is sometimes used in this place) reduces the level of intermodulation distortion. The emitter follower on transistor VT7 increases the current gain of the driver stage. Transistor VT9 serves to thermally stabilize the quiescent current of output transistors VT11, VT12 as their temperature increases. Increased output impedance is achieved using combined negative feedback (NOC) - voltage and current. The OOS voltage signal is removed from the output of the amplifier and is supplied through resistor R20 to the base of transistor VT4. The OOS current signal is removed from the resistor - current sensor R27 and is supplied to the base of the transistor VT4 through resistor R21. The somewhat unusual connection of the R9C4 circuit is used to eliminate noticeable DC voltage on the load due to current feedback. An experimental amplifier sample was tested to evaluate actual performance. An EMU0404 sound card and SpectraPLUS-SC software were used to measure distortion. Therefore, the measured distortion levels actually correspond to the sound card + amplifier system. In Fig. Figure 2 shows the frequency response of the total harmonic distortion of the amplifier.
Horizontally, it displays the frequency value of the test tone at which the distortion level was measured. The measurements used a mode with a DAC/ADC capacity of 24 bits and a sampling frequency of 192 kHz. The harmonics arising during the measurements were taken into account in a band up to 90 kHz, which is very important for the correct determination of the value of K at high frequencies. The increase in distortion at high frequencies is mainly due to a decrease in the depth of feedback with increasing frequency. The second of the main reasons is the increase in distortion of the input stage due to an increase in its output voltage, which is caused by a decrease in the gain of the stage on the transistor VT8. As can be seen, the harmonic coefficient is small even at high frequencies. In Fig. Figure 3 shows the spectrum of distortion at a frequency of 1 kHz.
As you can see, only the first three harmonics are present in it, the rest are below the measurement threshold. Such a narrow spectrum of distortion has a good effect on sound quality; as a result, the amplifier completely lacks “transistor sound”. In Fig. Figure 4 shows the spectrum of intermodulation distortion measured at frequencies of 18 and 19 kHz with an amplitude ratio of 1:1.
This is one of the most stringent tests that allows you to evaluate the linearity of an amplifier at high frequencies, where the depth of feedback is significantly reduced. The test allows you to identify non-linearity or poor high-frequency properties of the amplifier. As can be seen from Fig. 4, the amplitude of the difference frequency f 1 kHz is very small, which indicates the high linearity of the amplifier. The number of side frequencies that differ from the test ones by 1 kHz is also small. This suggests that the distortion spectrum remains narrow ("soft") even at high frequencies. All distortion measurements were carried out at an output power of 60 W into a 6 Ohm load when the amplifier was powered from a standard power supply. The measurement results show that in terms of distortion levels, this amplifier is not only not inferior to many expensive and famous industrial models, but also surpasses them. For a more clear comparison of the described amplifier with those presented in Fig. Figure 5 shows the dependence of the harmonic coefficient at a frequency of 1 kHz and a load of 4 Ohms on the output power of the UMZCH with a power supply designed for a power of 80 W in the load.
The output resistance (Rout) of the amplifier at the indicated values of the elements of the OOS circuits can be changed not only by choosing resistor R21, but also R27. The adjustment dependence on resistance R21 is shown in Fig. 6.
To obtain a higher output resistance, you should use the combined OOS calculation program on the editorial office's FTP server. If increasing this parameter is not required, then resistor R21 should be eliminated, and resistor R27 should be replaced with a wire jumper. Design and details. The amplifier is assembled on a printed circuit board, shown from the side of printed conductors in Fig. 7.
A resistor R12 is soldered on this side, intended for surface mounting of size 1208, but a resistor with axial leads can also be installed. In gray color in Fig. Figure 7 shows pieces of copper wire with a cross-section of 2.5 mm2, soldered onto a printed conductor to reduce its resistance. In Fig. Figure 8 shows the location of the housing elements.
Capacitor C12 is soldered to the terminals of resistor R20. If the amplifier is used in a stereo or multi-channel version, then it is advisable to use resistors included in the OOS circuit (R9, R20, R21), of high accuracy (deviation no more than ±1%) or select them with the same resistance for all channels. Resistors R24, R25, R27 - wirewound SQP-5 (SQP500JBR15,SQP-5W-R1 5-J) from YAGEO or made in China. Capacitors C2, SZ, C12 are ceramic with TKE group NPO, and C1, C7, C9, C10 are film capacitors for a voltage of at least 63 V. The ratings of all oxide capacitors correspond to the use of an amplifier in conjunction with a subwoofer.. If small-sized film capacitors are available, for example , from Epcos, then it is advisable to increase the capacitance of capacitors C7 and C10 to 1 µF. Oxide capacitors C5, C6, C8, C11 - any high-quality ones (with low equivalent series resistance). In position C4, you can also use a polar oxide capacitor, but you need to measure the polarity of the DC component at the output of the amplifier after assembly and resolder capacitor C4 in accordance with this polarity. During operation, the capacitors do not heat up, so it is more profitable to use capacitors with a permissible temperature of 85 ° C - their properties are slightly better. Complementary transistors 2N5551 and 2N5401 can be replaced with 2CS2240 and 2SA970, and 2SA1930 and 2SC5171 - with 2SA1358 and 2SC3421 or (which is somewhat worse ) on 2SB649 and 2SD669. Transistor VT9 - any p-p-p structure in an insulated TO-126 package. As outputs, you can use a pair of transistors IRFP240/IRFP9240. Power transistors are placed on heat sinks with an effective area of at least 700 cm2 each. They are insulated with mica gaskets or special heat-conducting films. To improve heat dissipation, it is necessary to use thermal conductive paste. An amplifier is a fairly high-frequency device, so to reduce possible interference from mobile communications, it is recommended to use ferrite rings on all cables (input, acoustic and power cables). The amplifier supply voltage is limited mainly by the permissible voltage of its semiconductor devices and capacitors and should not exceed +/-55 V. When installing capacitors in the power circuit (C5-C8, C10, C11) for an operating voltage of 80 V, the supply voltage can be increased to +/ -65 V. However, such an increase in the supply voltage is not recommended for working with a low-resistance load (4 Ohms). Setting up a properly assembled amplifier consists of setting the quiescent current of the output transistors with resistor R16 within 230. ..250 mA. After warming up at idle, the quiescent current must be adjusted. The quiescent current is determined by the voltage between the sources of the output transistors. An important role in the operation of the amplifier is played by its power supply. It also determines amplifier parameters such as maximum output power, overload capacity, background level and even the amount of distortion. The diagram of the amplifier power supply is shown in Fig. 9.
Capacitor C1 suppresses impulse noise coming from the network. Resistors R1 and R2 serve to discharge the filter capacitors when the power is turned off. The rectifier can use an integral diode bridge or individual diodes. Good results are obtained by using Schottky diodes. The maximum reverse voltage of the diodes must be at least 150-200 V, the maximum forward current depends on the output power of the amplifier and the number of its channels. For a subwoofer and stereo amplifier with an output power of no more than 80 W, the maximum forward current of the diodes should not be less than 10 A (for example, diode bridges RS1003-RS1007 or KVRS4002-KVRS4010). With a higher output power and/or a larger number of amplification channels, the rectifier diodes must be designed for a direct current of at least 20 A, for example, diode bridges KVRS4002-KVRS4010, KVRS5002-KVRS5010 or Schottky diodes 20CPQ150, 30CPQ150 with parallel connection of both diodes in the housing. In this case, it is recommended to increase the total capacitance of the filter capacitors to 30,000 µF per arm. To further reduce impulse noise coming from the network, each of the diodes can be shunted with a 0.01 μF capacitor to a voltage of at least 100 V. To select the required overall power of the transformer and the voltage on its secondary windings, depending on the required maximum output power of the amplifier, you can use the graphs in Fig. 10.
Black lines show graphs of the minimum power of the transformer. The solid line corresponds to the stereo amplifier; the dashed line corresponds to the subwoofer. The colored lines indicate the voltage on each of the secondary windings. It may seem strange that the transformer power of a stereo amplifier is less than twice its output power. Here there is a minimum transformer power in the input, sufficient for normal operation of the amplifier: the peak factor of audio signals is 12...16 dB, therefore the maximum output power of the amplifier is achieved relatively rarely and for a short time. This means that the average output power and current consumed from the power supply are several times less than the maximum. Therefore, the average power consumed from the transformer is several times less than the maximum. The transformer is designed for this average output power plus short-term peaks of maximum power, and with some margin. You can use a transformer with a larger overall power than shown in Fig. 10, but there is no point in exceeding this power more than twice. The amplifier does not contain a speaker system protection unit, so to protect it from direct voltage, you can use any of the designs described in the magazine or mentioned on this site.
Radio No. 10 2016 p. 8
Output stages based on "twos"
As a signal source we will use an alternating current generator with a tunable output resistance (from 100 Ohms to 10.1 kOhms) in steps of 2 kOhms (Fig. 3). Thus, when testing VCs at the maximum output resistance of the generator (10.1 kOhm), we will to some extent bring the operating mode of the tested VCs closer to a circuit with an open feedback loop, and in another (100 Ohm) - to a circuit with a closed feedback loop.
The main types of composite bipolar transistors (BTs) are shown in Fig. 4. Most often in VC, a composite Darlington transistor is used (Fig. 4a) based on two transistors of the same conductivity (Darlington “double”), less often - a composite Szyklai transistor (Fig. 4b) of two transistors of different conductivity with a current negative OS, and even less often - a composite Bryston transistor (Bryston, Fig. 4 c).
The "diamond" transistor, a type of Sziklai compound transistor, is shown in Fig. 4 g. Unlike the Szyklai transistor, in this transistor, thanks to the “current mirror”, the collector current of both transistors VT 2 and VT 3 is almost the same. Sometimes the Shiklai transistor is used with a transmission coefficient greater than 1 (Fig. 4 d). In this case, K P =1+ R 2/ R 1. Similar circuits can be obtained using field-effect transistors (FETs).
1.1. Output stages based on "twos". "Dvoyka" is a push-pull output stage with transistors connected according to the Darlington, Sziklai circuit or their combination (quasi-complementary stage, Bryston, etc.). A typical push-pull output stage based on a Darlington deuce is shown in Fig. 5. If emitter resistors R3, R4 (Fig. 10) of input transistors VT 1, VT 2 are connected to opposite power buses, then these transistors will operate without current cut-off, i.e. in class A mode.
Let's see what pairing the output transistors will give for the two "Darlingt she" (Fig. 13).
In Fig. Figure 15 shows a VK circuit used in one of the professional and onal amplifiers.
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The Siklai scheme is less popular in VK (Fig. 18). At the early stages of the development of circuit design for transistor UMZCHs, quasi-complementary output stages were popular, when the upper arm was performed according to the Darlington circuit, and the lower one according to the Sziklai circuit. However, in the original version, the input impedance of the VC arms is asymmetrical, which leads to additional distortion. A modified version of such a VC with a Baxandall diode, which uses the base-emitter junction of the VT 3 transistor, is shown in Fig. 20.
In addition to the considered “twos,” there is a modification of the Bryston VC, in which the input transistors control transistors of one conductivity with the emitter current, and the collector current controls transistors of a different conductivity (Fig. 22). A similar cascade can be implemented on field-effect transistors, for example, Lateral MOSFET (Fig. 24).
The hybrid output stage according to the Sziklai circuit with field-effect transistors as outputs is shown in Fig. 28. Let's consider the circuit of a parallel amplifier using field-effect transistors (Fig. 30).
As an effective way to increase and stabilize the input resistance of a “two”, it is proposed to use a buffer at its input, for example, an emitter follower with a current generator in the emitter circuit (Fig. 32).
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Of the “twos” considered, the worst in terms of phase deviation and bandwidth was the Szyklai VK. Let's see what using a buffer can do for such a cascade. If instead of one buffer you use two on transistors of different conductivities connected in parallel (Fig. 35), then you can expect further improvement in parameters and an increase in input resistance. Of all the considered two-stage circuits, the Szyklai circuit with field-effect transistors showed itself to be the best in terms of nonlinear distortions. Let's see what installing a parallel buffer at its input will do (Fig. 37).
The parameters of the studied output stages are summarized in Table. 1.
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Analysis of the table allows us to draw the following conclusions:
- any VC from the “twos” on the BT as a UN load is poorly suited for work in a high-fidelity UMZCH;
- the characteristics of a VC with a DC at the output depend little on the resistance of the signal source;
- a buffer stage at the input of any of the “twos” on the BT increases the input impedance, reduces the inductive component of the output, expands the bandwidth and makes the parameters independent of the output impedance of the signal source;
- VK Siklai with a DC output and a parallel buffer at the input (Fig. 37) has the highest characteristics (minimum distortion, maximum bandwidth, zero phase deviation in the audio range).
Output stages based on "triples"
In high-quality UMZCHs, three-stage structures are more often used: Darlington triplets, Shiklai with Darlington output transistors, Shiklai with Bryston output transistors and other combinations. One of the most popular output stages at present is a VC based on a composite Darlington transistor of three transistors (Fig. 39). In Fig. Figure 41 shows a VC with cascade branching: the input repeaters simultaneously operate on two stages, which, in turn, also operate on two stages each, and the third stage is connected to the common output. As a result, quad transistors operate at the output of such a VC.
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The VC circuit, in which composite Darlington transistors are used as output transistors, is shown in Fig. 43. The parameters of the VC in Fig. 43 can be significantly improved if you include at its input a parallel buffer cascade that has proven itself well with “twos” (Fig. 44).
Variant of VK Siklai according to the diagram in Fig. 4 g using composite Bryston transistors is shown in Fig. 46. In Fig. Figure 48 shows a variant of the VK on Sziklai transistors (Fig. 4e) with a transmission coefficient of about 5, in which the input transistors operate in class A (thermostat circuits are not shown).
In Fig. Figure 51 shows the VC according to the structure of the previous circuit with only a unit transmission coefficient. The review will be incomplete if we do not dwell on the output stage circuit with Hawksford nonlinearity correction, shown in Fig. 53. Transistors VT 5 and VT 6 are composite Darlington transistors.
Let's replace the output transistors with field-effect transistors of the Lateral type (Fig. 57
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Anti-saturation circuits of output transistors contribute to increasing the reliability of amplifiers by eliminating through currents, which are especially dangerous when clipping high-frequency signals. Variants of such solutions are shown in Fig. 58. Through the upper diodes, excess base current is discharged into the collector of the transistor when approaching the saturation voltage. The saturation voltage of power transistors is usually in the range of 0.5...1.5 V, which approximately coincides with the voltage drop across the base-emitter junction. In the first option (Fig. 58 a), due to the additional diode in the base circuit, the emitter-collector voltage does not reach the saturation voltage by about 0.6 V (voltage drop across the diode). The second circuit (Fig. 58b) requires the selection of resistors R 1 and R 2. The lower diodes in the circuits are designed to quickly turn off the transistors during pulse signals. Similar solutions are used in power switches.
Often, to improve the quality, UMZCHs are equipped with separate power supply, increased by 10...15 V for the input stage and voltage amplifier and decreased for the output stage. In this case, in order to avoid failure of the output transistors and reduce the overload of the pre-output transistors, it is necessary to use protective diodes. Let's consider this option using the example of modification of the circuit in Fig. 39. If the input voltage increases above the supply voltage of the output transistors, additional diodes VD 1, VD 2 open (Fig. 59), and the excess base current of transistors VT 1, VT 2 is dumped onto the power buses of the final transistors. In this case, the input voltage is not allowed to increase above the supply levels for the output stage of the VC and the collector current of transistors VT 1, VT 2 is reduced.
Bias circuits
Previously, for the purpose of simplicity, instead of a bias circuit in the UMZCH, a separate voltage source was used. Many of the considered circuits, in particular, output stages with a parallel follower at the input, do not require bias circuits, which is their additional advantage. Now let's look at typical displacement schemes, which are shown in Fig. 60, 61.
Stable current generators. A number of standard circuits are widely used in modern UMZCHs: a differential cascade (DC), a current reflector ("current mirror"), a level shift circuit, a cascode (with serial and parallel power supply, the latter is also called a "broken cascode"), a stable generator current (GST), etc. Their correct use can significantly improve the technical characteristics of UMZCH. We will estimate the parameters of the main GTS circuits (Fig. 62 - 6 6) using modeling. We will assume that the GTS is a load of the UN and is connected in parallel with the VC. We study its properties using a technique similar to the study of VC.
Current reflectors
The considered GST circuits are a variant of a dynamic load for a single-cycle UN. In an UMZCH with one differential cascade (DC), to organize a counter dynamic load in the UN, they use the structure of a “current mirror” or, as it is also called, a “current reflector” (OT). This structure of the UMZCH was characteristic of the amplifiers of Holton, Hafler, and others. The main circuits of the current reflectors are shown in Fig. 67. They can be either with a unity transmission coefficient (more precisely, close to 1), or with a greater or lesser unit (scale current reflectors). In a voltage amplifier, the OT current is in the range of 3...20 mA: Therefore, we will test all OTs at a current of, for example, about 10 mA according to the diagram in Fig. 68.
The test results are given in table. 3.
As an example of a real amplifier, the S. BOCK power amplifier circuit, published in the journal Radiomir, 201 1, No. 1, p. 5 - 7; No. 2, p. 5 - 7 Radiotechnika No. 11, 12/06
The author's goal was to build a power amplifier suitable for both sounding "space" during festive events and for discos. Of course, I wanted it to fit in a relatively small-sized case and be easily transported. Another requirement for it is the easy availability of components. In an effort to achieve Hi-Fi quality, I chose a complementary-symmetrical output stage circuit. The maximum output power of the amplifier was set at 300 W (into a 4 ohm load). With this power, the output voltage is approximately 35 V. Therefore, the UMZCH requires a bipolar supply voltage within 2x60 V. The amplifier circuit is shown in Fig. 1. The UMZCH has an asymmetrical input. The input stage is formed by two differential amplifiers.
A. PETROV, Radiomir, 201 1, No. 4 - 12
The writing of this material was prompted by an article in, in which the author strongly criticizes everything that has been done so far in the field of development of audio frequency amplifiers, and proposes the structure of his “absolute” UMZCH. I do not agree with the author, and therefore, based on the analysis of known developments on individual UMZCH units, I want to present my version of a simple, “flawless”, in the words of Douglas Self, UMZCH.
Today, three main disadvantages of bipolar transistors are known:
Early effect (dependence of collector current on emitter-collector voltage);
Miller effect (dependence of input capacitance on gain);
Thermal distortions associated with the dependence of parameters on crystal temperature.
A generally accepted way to combat the first two disadvantages and partly the third is through cascode circuits. The reduction of the first effect, which is also associated with power supply pulsations of the UMZCH under load, is facilitated by separate power supply of the driver and the output stage. To eliminate thermal distortion, it is necessary to stabilize the power dissipated by the transistor, and if this is not doneperhaps, then at least minimize its fluctuations.
First, let's decide on the driver. As studies in, and later in, showed, extremely simple symmetrical cascode drivers are not inferior, and in some cases surpass the parameters of more complex circuits using a differential cascade (DC). Therefore, we will focus on the cascode driver.
Now you need to select the output stage (VC). The simplest option, not much inferior to the modified Hawksford VC, is the economical Shikpai VC with composite Darlington transistors, at the input of which a parallel is addedny repeater. In this VC, the base-emitter junctions of the parallel follower set the bias for the VC and simultaneously carry out thermal stabilization. To do this you need to select transistors VT 12, VT 16 and VT 13, VT 1 5 of the same type and in pairs to ensure thermal contact.
The advantage of this solution is that these transistors work as a current mirror, and by changing the collector current of the parallel follower transistors, you can adjust the quiescent current of the output transistors. In such a connection, distortion depends little on the quiescent current of the output transistors, therefore, in order to increase efficiency, it can be set within 5...30 mA. Another advantage of this VC is that it introduces very little distortion even without OOS.
Diodes VD 5, VD 8 improve thermal stabilization and reduce distortion, since the output transistors act as large-scale current reflectors with a high reflection coefficient, and diodes VD 6, VD 7 serve to limit the minimum base-collector voltage of the output transistors in order to prevent their saturation. Low resistance resistors R 29, R 30 promote rapid switching off of transistors.
As a result of combining these two cascades, we obtain a UMZCH circuit with a single-stage driver, shown in Fig. 1.
The advantage of a completely symmetrical UMZCH circuit is that when selecting “mirror” transistors according to the static transfer coefficient of the base current (for yourself, your loved one, you can afford this) and identical electrolytic capacitors, the UMZCH has no transient processes. Therefore, there is no need for a delay relay for connecting the speakers.
In order to minimize distortions associated with the listed shortcomings, a slight complication of the driver circuit has been made: a helmet has been added A d for input transistors and as a stable generatorcurrent (GTS) used Douglas Self's favorite GTS withcurrent feedback system, in which the collector currents of the current feedback transistors are stabilized. Such a GTS makes it possible to minimize the influence of supply voltage pulsations and, thus, eliminate the need for additional power sources. The most linear section of the stabilization current characteristic for diode E202(S 202) - when the voltage drop across it is within 5...20 (3...50) V. The diode drop is limited taking into account the voltage drop under load using a resistor R 18. If there is no diode, it can be replaced with a jumper; this will hardly affect the parameters.
Old-style transistors such as KT825, KT827 (analogs of those shown in the diagram) can be successfully used as output transistors. Even better results are possibleget with modern transistors, for example, 2SD 2560,2SB 1647; 2SD 2449, 2SB 1594; 2SD 2385, 2SB 1556 and similar.
The zero offset at the output of the UMZCH is processed by the integrator at DA 1. Thanks to additional filtering, it does not manifest itself in any way in the audio range. Considering that the used VC itself has low distortion, it is possible to provide jumpers for operation without general OOS, as proposed in.
This amplifier has an open input, so before connecting a normalizing amplifier to it, you must make sure that there is no DC component at its output. The input resistance of the UMZCH is small (about 3 kOhm), so if there is a capacitor at the output of the normalizing amplifier, its capacitance must be at least 10 μF. Becausenon-electrolytic capacitors of such capacity are large enough; you can make a capacitor from two back-to-back polar ones with a capacity of 22...47 μF and a non-polar capacitor with a capacity of 1...2 μF in parallel. It is better to use a buffer repeater after the volume control (andif the sensitivity is not enough, then a normalizing amplifier with K and = 2...3) to the op-amp and connect the UMZCH directly to its output.
Let's take standard characteristics: a Bode diagram without capacitor C1, nonlinear distortions at frequencies of 1, 10 and 20 kHz, and also see if there are visible distortions in the signal shape at a frequency of 100 kHz.
The Bode diagram is shown in Fig. 2. It shows that the amplifier is quite wideband: the cutoff frequency is about 500 kHz with a unity gain frequency of 2 MHz. SmallThe surge in the 400 kHz region is due to the operation of bipolar correction. The amplitude margin is 18 dB, the phase margin is about 60°, which is the optimal value.
The introduced nonlinear distortions at an output signal amplitude of 30 V at frequencies of 1.10 and 20 kHz are respectively equal to 0.0005, 0.001 and 0.003%. As an example, Fig. 3 shows the distortion spectrum at a frequency of 10 kHz.
As can be seen from the figure, the spectrum contains only the 2nd and 3rd harmonics. The level of the nearest harmonic falling within the audio range is the same 0.0005% as at a frequency of 1 kHz.
Let's check the slew rate of the signal: is there any visible distortion at full power at a frequency of 100 kHz (Fig. 4)?
As we see, and everything is fine here. When checking the UMZCH with a meander frequency of 2 kHz(without capacitor C1) it turned out that small emissions were observed on the shelves at the end of the front. But with the installation of capacitor C1 in place, the meander shelves are absolutely flat, and the signal edges are quite steep.
The second modification of the UMZCH, which I also want to pay attention to, is shown in Fig. 5. The number of elements in it is the same as in the circuit in Fig. 1, but the output stage of the driver, like the input stage, is cascode.