Electromagnetic interference suppression filters from Epcos. Example of Noise Suppression in AC Power Supplies

Pulse noise suppressor for P399A.

Over the past few months, when the street lighting was turned on, it became almost impossible for me to work on the air due to the presence of strong interference from DRL lamps. My device is not imported, but a transceiver P399A, which is used as a base unit for VHF (“Hyacinth” is used as a reference oscillator in HF synthesizers for set-top boxes). Having gone on vacation, I decided to somehow deal with the problem that had arisen, and within a week the proposed “Pulse Interference Suppressor (PIP)” was designed.

The schematic diagram of the device is shown in Fig. 1. The PIP consists of two units: a peak detector and a pulse suppression unit. The device is turned on between the second mixer and the amplifier (path 215 kHz).

The peak detector circuit with some modifications was borrowed from the magazine “Ham Radio, 2, 1973, W2EGH”, in particular, chains D1, R6, S1 and D2, R7, S2 were added, and the suppressor unit was made according to the controlled attenuator circuit R16, C18, Q4, the introduction of which, among other things, somewhat improved dynamic range Receiver AGC. The use of LC delay lines common for these devices did not provide any identified advantage. Probably due to their narrowband due to low IF and as a result of the “stretching” of the interference pulse. Application at the input of a peak detector broadband amplifier on the KT610A transistor is due to the need to obtain an undistorted output signal with an amplitude of up to 20V and, accordingly, minimal impact on the duration and shape of the original interference pulse. The use of additional AGC in the amplifier only worsened its operation, but the introduction of the D2, R7 chain automatically blocks the operation of the PIP in the presence of a powerful useful signal (tested up to +60 dB according to real signal from the air at full gain R1). S1 – “Deep suppression” allows you to eliminate even small interference only with very low levels useful signal (tested when receiving EME stations in JT65B mode), when the signal strength is S2 or more, the detected envelope is superimposed on the signal. The quality of decoding in FSK441 mode has not yet been really tested.

The PIP scheme is still in the process of being finalized, but, nevertheless, it can already provide a good service for real work on the air to those who need it. Any modification and publication that improves the parameters of the device is also welcome.

In switching power supplies, noise occurs during switching key elements. This interference is induced by the power cable connected to the network AC. Therefore, measures must be taken to suppress them.

Typical solution for an electromagnetic interference network filter for a switching power supply

To suppress interference penetrating through the power cable into the primary circuit from pulse source power supply, the circuit shown in Figure 9 is used.

Figure 9 - Suppression of interference penetrating through the cable

Differential and common mode interference

There are two types of interference: differential and common mode. The differential noise current induced on both wires of the power line flows through them in opposite directions, as shown in Figure 10. The common mode noise current flows through all lines in the same direction, see Figure 11.

Figure 10 - Differential interference

Figure 11 - Common mode interference

Functional purpose of network filter elements

The figures below provide examples of the use of various filter elements and graphics that illustrate the effect of their use. The given graphs show the change in the intensity of differential and common-mode noise of a switching power supply relative to the level of industrial noise. Figure 12 shows signal graphs in the absence of a filter at the input of a switching power supply. As can be seen from the graph, the level of differential and common-mode interference is quite high. Figure 13 illustrates an example of using an X filter capacitor. The graph shows a noticeable decrease in the level of differential interference.

Figure 14 shows the results sharing X-capacitors and Y-capacitors. The graph clearly shows the effective suppression of both common-mode and differential-mode interference. The use of X-capacitors and Y-capacitors in combination with a common mode choke (common mode choke) is shown in Figure 15. The graph shows a further reduction in the level of both differential and common mode noise. This is because a real common mode choke has some differential inductance.

Figure 12 - Without filter

Figure 13 - Using an X-capacitor

Figure 14 - Using X-capacitor and Y-capacitor

Figure 15 - Using X-capacitor, Y-capacitor and common mode choke

Example of interference suppression in a mobile phone

Radiated Interference Sources

The interference generated by the signal processing block passes into the RF block, which leads to a significant deterioration in sensitivity. Signal processing block mobile phone, which is typically built around a baseband signal processing IC, controls various signals such as speech signal and LCD signal. The signal processing IC is a source of significant noise because it operates on high frequency and a plurality of data lines are connected to it. When noise passes through the data lines or power/GND buses from the signal processing unit to the RF unit, its sensitivity deteriorates, resulting in an increase in the Bit Error Rate (BER).

Components for interference suppression in mobile phones

To improve the BER (Bit Error Rate) parameter, that is, reduce the percentage of received erroneous bits, it is necessary to suppress interference penetrating from the signal processing block to the RF block. To do this, install EMI filters on all buses connecting these units. In addition, it is also important to shield the signal processing unit, since the noise level it emits is latest models mobile phones has increased significantly.

Installing Filters on the Display Control Bus

The LCD control bus contains many signal lines switching simultaneously, causing a significant increase in pulse current flowing in the ground (GND) and power circuits. Therefore, it is necessary to limit the current flowing through the signal lines. Typically, BLA31 series ferrite chip bead arrays and NFA31G series EMIFIL® chip filters with resistor are used for this purpose. If for design reasons the use of these components is not possible, then EA series EMC absorbers should be used to suppress interference passing through the LCD display flex cable.

Improved shielding

Typically, a conductive coating is applied to the inner surface of the plastic case of a mobile phone. As the functionality of a mobile phone increases, the level of interference from the signal processing unit also increases. Therefore, it is necessary to shield the signal processing unit with the same care as the RF unit. When designing a mobile phone case, to reduce impedance at high frequencies, you should try to ensure as large a contact area as possible between the parts of the case. To improve shielding, metal shielding elements or EMC absorbers should be used in the signal processing unit, where possible.

Shevkoplyas B.V. “Microprocessor structures. Engineering solutions." Moscow, publishing house "Radio", 1990. Chapter 4

4.1. Suppression of interference via the primary supply network

Waveform AC voltage industrial power supply network (~"220 V, 50 Hz) for short periods of time can differ greatly from a sinusoidal one - surges or “insertions” are possible, a decrease in the amplitude of one or several half-waves, etc. The reasons for the occurrence of such distortions are usually associated with a sharp change network load, for example when turning on a powerful electric motor, oven, welding machine. Therefore, whenever possible, isolation from such sources of interference should be carried out via the network (Fig. 4.1).

Rice. 4.1 Connection options digital device to the primary power supply

In addition to this measure, it may be necessary to introduce surge protector at the device power input in order to suppress short-term interference. The resonant frequency of the filter can be in the range of 0.1.5-300 MHz; broadband filters provide interference suppression over the entire specified range.

Figure 4.2 shows an example of a network filter circuit. This filter has dimensions of 30 X30x20 mm and is mounted directly on the network input block to the device. Filters must use high frequency capacitors and inductors, either coreless or with high frequency cores.

In some cases, it is necessary to introduce an electrostatic shield (a regular water pipe connected to a grounded power panel housing) to lay the primary power supply wires inside it. As noted in, a short-wave transmitter of a taxi fleet, located on the opposite side of the street, is capable of transmitting signals with an amplitude of several hundred volts on a piece of wire at a certain relative orientation. The same wire, placed in an electrostatic shield, will be reliably protected from this kind of interference.

Rice. 4.2. Example of a network filter circuit

Let's look at methods for suppressing network interference directly in the device's power supply. If the primary and secondary windings power transformer located on the same coil (Fig. 4.3, a), then due to capacitive coupling between the windings, impulse noise can pass from the primary circuit to the secondary. Four methods for suppressing such interference are recommended (in order of increasing effectiveness).

1. The primary and secondary windings of the power transformer are made on different coils (Fig. 4.3, b). The throughput capacitance C decreases, but the efficiency decreases, since not all of the magnetic flux from the primary winding area enters the secondary winding area due to scattering through the surrounding space.
2. The primary and secondary windings are made on the same coil, but are separated by a copper foil screen with a thickness of at least 0.2 mm. The screen should not be a short-circuited loop. It is connected to the body ground of the device (Fig. 4.3, c)
3. The primary winding is completely enclosed in a screen that is not a short-circuited turn. The screen is grounded (Fig. 4.3, G).
4. The primary and secondary windings are enclosed in individual screens, between which a separating screen is laid. The entire transformer is enclosed in a metal casing (Fig. 4.3,<Э). Экраны и корпус заземляются. Этот тип трансформатора в силу предельной защищенности от прохождения помех получил название «ультраизолятор».

With all of the listed methods of noise suppression, the wiring of network wires inside the device should be done using a shielded wire, connecting the screen to the chassis ground. Invalid uk
wiring into one bundle of network and other (power boards, signal, etc.) wires" even in the case of shielding of both.

It is recommended to install a capacitor with a capacity of approximately 0.1 μF parallel to the primary winding of the power transformer in close proximity to the winding terminals and a current-limiting resistor with a resistance of about 100 ohms in series with it. This allows the energy stored in the power transformer core to be “shorted” at the moment the mains switch opens.

Rice. 4.3. Options for protecting a power transformer from the transmission of impulse noise from the network to the secondary circuit (and vice versa):
a—no protection; b - separation of the primary and secondary windings; V- laying the screen between the windings; G - complete shielding of the primary winding; d — complete shielding of all transformer elements

Rice. 4.4. Simplified power supply diagram (A) and diagrams (b, c), explaining the operation of a full-wave rectifier.

The power supply is the greater source of impulse noise on the network, the larger the capacitance of capacitor C

Note that with an increase in the capacitance C of the filter (Fig. 4.4, a) of the power supply of our device, the probability of failures of neighboring devices increases, since the energy consumption from the network by our device increasingly takes on the nature of shocks. Indeed, the voltage at the output of the rectifier also increases during those time intervals when energy is taken from the network (Fig. 4.4, b). These intervals in Fig. 4.4 are shaded.

With increasing capacitance of the capacitor C, the periods of its charge become shorter and shorter (Fig. 4.4, c), and the current taken in a pulse from the network becomes larger. Thus, an apparently “harmless” device can create interference in the network that is “not inferior” to the interference from a welding machine.

4.2. Grounding rules providing protection against ground interference

In devices made in the form of structurally complete blocks, there are at least two types of ground buses—case and circuit. According to safety requirements, the housing bus must be connected to the grounding bus laid in the room. The circuit bus (relative to which the signal voltage levels are measured) should not be connected to the chassis bus inside the unit; a separate terminal isolated from the chassis should be provided for it.

Rice. 4.5. Improper and correct grounding of digital devices. Shown is the ground bus that is usually present indoors.

In Fig. Figure 4.5 shows options for incorrect and correct grounding of a group of devices that are interconnected by information lines. (these lines are not shown). Circuit ground buses are connected by individual wires at point A, and case buses are connected at point B, as close as possible to point A. Point A may not be connected to the ground bus in the premises, but this creates inconvenience, for example, when working with an oscilloscope, which The probe's ground is connected to the housing.

If the grounding is incorrect (see Fig. 4.5), the pulse voltages generated by equalizing currents along the ground bus will actually be applied to the inputs of the receiving main elements, which can cause their false operation. It should be noted that the choice of the best grounding option depends on specific “local” conditions and is often carried out after a series of careful experiments. However, the general rule (see Figure 4.5) always remains in force.

4.3. Suppression of interference in secondary power supply circuits

Due to the finite inductance of the power and ground buses, pulsed currents cause the appearance of pulsed voltages of both positive and negative polarity, which are applied between the power and ground pins of the microcircuits. If the power and ground buses are made of thin printed or other conductors, and high-frequency decoupling capacitors are either completely absent or their number is insufficient, then when several TTL microcircuits are simultaneously switched at the “far” end of the printed circuit board, the amplitude of power supply pulse noise (voltage surges acting between the power pin and the ground of the microcircuit) can be 2 V or more. Therefore, when designing a printed circuit board, the following recommendations must be followed.

1. The power and ground buses must have minimal inductance. To do this, they are made in the form of lattice structures covering the entire area of ​​the printed circuit board. It is unacceptable to connect TTL microcircuits to a bus that is a “tap”, since as it approaches its end, the inductance of the power supply circuits accumulates. The power and ground buses should, if possible, cover the entire free area of ​​the printed circuit board. Particular attention should be paid to the design of dynamic memory storage matrices on K565RU5, RU7, etc. chips. The matrix should be a square so that the address and control lines have a minimum length. Each microcircuit must be located in an individual cell of a lattice structure formed by power and ground buses (two independent grids). The power and ground buses of the storage matrix should not be loaded with “foreign” currents flowing from address drivers, control signal amplifiers, etc.
2. Connecting external power and ground buses to the board through a connector must be done through several contacts evenly spaced along the length of the connector, so that the power and ground bus lattice structures can be entered from several points at once.
3. Suppression of power supply interference should be carried out close to where it occurs. Therefore, a high-frequency capacitor with a capacity of at least 0.02 μF must be located near the power pins of each TTL chip. This also applies particularly to the mentioned dynamic memory chips. To filter low-frequency noise, it is necessary to use electrolytic capacitors, for example, with a capacity of 100 μF. When using dynamic memory chips, electrolytic capacitors are installed, for example, in the corners of the storage matrix or in another place, but close to these chips.

Accordingly, instead of high-frequency capacitors, special power buses BUS-BAR, CAP-BUS are used, which are laid under the lines of microcircuits or between them, without disturbing the usual automated technology for installing elements on the board with subsequent wave soldering. These buses are distributed capacitors with a linear capacitance of approximately 0.02 μF/cm. For the same total capacitance as discrete capacitors, the busbars provide significantly better noise rejection at higher packing densities.

Rice. 4.6. Options for connecting P1-PZ boards to the power supply

In Fig. 4.6 provides recommendations for connecting devices made on printed circuit boards P1-PZ to the output of the power supply. A high-current device made on a PZ board creates more noise on the power and ground buses, so it should be physically closer to the power supply, or even better, provide its power using individual buses.

4.4. Rules for working with agreed communication lines

In Fig. Figure 4.7 shows the shape of the signals transmitted along the cable, depending on the ratio of the resistance of the load resistor R and the characteristic impedance of the cable p. Signals are transmitted without distortion at R=p. The characteristic impedance of a particular type of coaxial cable is known (for example, 50, 75, 100 ohms). The characteristic impedance of flat cables and twisted pairs is usually close to 110-130 Ohms; its exact value can be obtained experimentally by selecting a resistor K, when connected, the distortion is minimal (see Fig. 4.7). When conducting an experiment, you should not use variable resistance wires, as they have a high inductance and can distort the signal shape.

Communication line of the “open collector” type (Fig. 4.8). To transmit each main signal with a rise time of about 10 ns at distances exceeding 30 cm, a separate twisted pair is used or one pair of cores is allocated in a flat cable. In the passive state, all transmitters are turned off. When any transmitter or group of transmitters is triggered, the line voltage drops from above 3 V to approximately 0.4 V.

With a line length of 15 m and with proper matching, the duration of transient processes in it does not exceed 75 ns. The line implements the OR function with respect to signals represented by low voltage levels.

Rice. 4.7. Transmission of signals via cable. O—voltage pulse generator

Communication line of the “open emitter” type (Fig. 4.9"). This example shows a line option using a flat cable. Signal wires alternate with ground wires. Ideally, each signal wire is bordered on both sides by its own ground wires, but this, as a rule, is not particularly necessary. In Fig. 4.9, each signal wire is adjacent to “own” and “foreign” ground, which is usually quite acceptable. A flat cable and a set of twisted pairs are essentially almost the same thing, and yet the second is preferable in conditions of increased levels of external interference. The open emitter line implements the OR function with respect to signals represented by high voltage levels. The timing characteristics approximately correspond to those of an “open collector” line.

Communication line of the “differential pair” type (Fig. 4.10). The line is used for unidirectional signal transmission and is characterized by increased noise immunity, since the receiver reacts to the difference in signals, and externally induced interference affects both wires approximately equally. The length of the line is practically limited by the ohmic resistance of the wires and can reach several hundred meters.

Fig, 4.8. Open collector communication line

Rice. 4.9. Open emitter communication line

Rice. 4.10. Differential pair communication line

All lines considered should use receivers with high input impedance, low input capacitance, and preferably with a hysteretic transfer characteristic to increase noise immunity.

Physical implementation of the highway (Fig. 4. II), Each device connected to the trunk contains two connectors. A diagram similar to that shown in Fig. 4.11 was discussed earlier (see Fig. 3.3), so we will focus only on the rules that must be followed when designing matching blocks (MBs).

Transmission of main signals through connectors. The best options for wiring connectors are shown in Fig. .4.12. In these cases, the front of the pulse traveling along the main line almost “does not feel” the connector, since the heterogeneity introduced into the cable line is insignificant. In this case, however, it is required to occupy 50% of the used contacts underground.

If for some reason this condition cannot be met, then, at the expense of noise immunity, you can adopt a second, more economical option for the number of contacts for wiring the connectors, shown in Fig. 4.13. This option is often used in practice. Twisted pair grounds (or flat cable grounds) are assembled onto metal strips with a cross-section that is as large as possible, for example 5 mm2.

The wiring of these lands is carried out evenly along the length of the strip, as the corresponding signal wires are soldered. Both strips are combined through a connector using a series of jumpers of minimum length and maximum cross-section, and the jumpers are located evenly along the length of the strips. Each ground jumper should correspond to no more than four signal lines, but the total number of jumpers should not be less than three (one in the center and two at the edges).

Rice. 4.13. Acceptable option for transmitting signals through the connector. Н-=5 mm2—bar cross-section, 5^0.5 mm2—ground wire cross-section

Rice. 4.14. Options for making branches from the main line

Making branches from the main line. In Fig. Figure 4.14 shows options for incorrect and correct execution of a branch from the main line. The path of one line is traced, the ground wire is shown conditionally. The first option (a typical mistake of novice circuit designers!) is characterized by splitting the wave energy into two parts,

Rice. 4.15. Options for connecting receivers to the highway
coming from line A. One part goes to the charge of line B, the other to the charge of line C. After the charge of line C, the “full” wave begins to propagate along line B, trying to catch up with the previously departed wave with half the energy. The signal front thus has a stepped shape.

If the branch is performed correctly, the segments of lines A, C and B are connected in series, so the wave practically does not split and the signal fronts are not distorted. Transmitters and receivers located on the board should be as close as possible to its edge to reduce the heterogeneity introduced at the point where lines B and C join together.

To decouple receiver bundles from the backbone, you can use one or bidirectional transceivers (see Fig. 3.18, 3.19). When branching a line into several directions, a separate transmitter should be allocated for each (Fig. 4.15, V).

For transmission over a line, it is better to use trapezoidal rather than rectangular pulses. Signals with flat fronts, as noted, propagate along the line with less distortion. In principle, in the absence of external interference, for any arbitrarily long and even unmatched line, it is possible to select such a slow signal rise rate that the transmitted and received signals will differ by an arbitrarily small amount.

To receive trapezoidal pulses, the transmitter is designed as a differential amplifier with an integrating feedback circuit. At the input of the main receiver, also made in the form of a differential amplifier, an integrating circuit is installed to filter high-frequency interference.

When transmitting signals within a board, when the number of receivers is large, “serial matching” is often used. It consists in the fact that a resistor with a resistance of 20-50 Ohms is connected in series with the output of the transmitter, in the immediate vicinity of this output. This allows you to suppress oscillatory processes at the signal fronts. This technique is often used when transmitting control signals (KA5, SAZ, \UE) from amplifiers to LSI dynamic memory.

4.5. About the protective properties of cables

In Fig. 4.16a shows the simplest scheme for transmitting signals over a coaxial cable, which in some cases can be considered quite satisfactory. Its main disadvantage is that in the presence of pulse equalizing currents between the frame grounds (potential equalization is the main function of the frame ground system), part of these currents 1 can flow along the cable braid and cause a voltage drop (mainly due to the inductance of the braid), which ultimately acts on the load K.

Moreover, in this sense, the diagram shown in Fig. 4.16, a, turns out to be preferable, and with an increase in the number of points of contact between the cable braid and the body ground, the possibilities for the drainage of induced charges from the braid improve. The use of a cable with additional braiding (Fig. 4.16, c) makes it possible to protect against both capacitive interference and equalizing currents, which in this case flow through the outer braid and have virtually no effect on the signal circuit.

Connecting a cable with additional braiding according to the diagram shown in Fig. 4.16, d, allows you to improve the frequency properties of the line by reducing its linear capacitance. In an ideal case, the potential of any elementary section of the central core coincides with the potential of the elementary cylinder of the inner braid surrounding this section.

Lines of this type are used in local computer networks to increase the speed of information transfer. The outer braid of the cable is part of the signal circuit, and therefore this circuit is equivalent in terms of immunity to external interference to the circuit shown in Fig. 4.16.6.

Rice. 4.16. Cable options

Neither the copper nor the aluminum braiding of a simple coaxial cable protects it from low-frequency magnetic fields. These fields induce an emf both on the section of the braid and on the corresponding section of the central core.

Although these EMFs are of the same sign, they do not compensate each other in magnitude due to the different geometries of the corresponding conductors - the central core and the braid. The differential emf is ultimately applied to the load K. Additional braid (Fig. 4. 16, c, d) is also unable to prevent the penetration of a low-frequency magnetic field into its internal region

Protection from low-frequency magnetic fields is provided by a cable containing a twisted pair of wires enclosed in a braid (Fig. 4.16, d). In this case, the EMF induced by an external magnetic field on the wires that make up the twisted pair completely compensate each other both in sign and in absolute value.

This is all the more true the smaller the wire twisting step is compared to the field action area and the more carefully (symmetrically) the twisting is performed. The disadvantage of such a line is its relatively low frequency “ceiling”—about 15 MHz—due to large energy losses of the useful signal at higher frequencies.

The diagram shown in Fig. 4.16, e, provides the best protection against all types of interference (capacitive interference, equalizing currents, low-frequency magnetic fields, high-frequency electromagnetic fields).

It is recommended to connect the inner braid to the “radio” or “true” (literally grounded) ground, and the outer braid to the “system” (circuit or case) ground. In the absence of a “true” ground, you can use the connection circuit shown in Fig. 4. 16, and.

The outer braid connects to the system ground at both ends, while the inner braid connects only to the source side. In cases where there is no need for protection from low-frequency magnetic fields and it is possible to transmit information without using paraphase signals, one of the twisted pair wires can serve as a signal wire, and the second as a screen. In these cases, the diagrams shown in Fig. 4.16, c,f, can be thought of as coaxial cables with three shields - the twisted pair ground wire, the inner and outer braids of the cable.

4.6. Using optocouplers to suppress interference

If the system devices are separated by a considerable distance, for example 500 m, then it is difficult to count on the fact that their lands always have the same potential. As noted, equalizing currents through ground conductors create impulse noise on these conductors due to their inductance. This noise is ultimately applied to the receivers' inputs and can cause false positives.

The use of lines of the “differential pair” type (see § 4.4) allows you to suppress only common-mode interference and therefore does not always produce positive results. In Fig. Figure 4.17 shows diagrams of optocoupler isolation between two devices remote from each other.

Rice. 4.17. Schemes of optocoupler isolation between devices remote from each other:
a - with an active receiver, b- with active transmitter

The circuit with an “active receiver” (Fig. 4.17, a) contains a transmitting optocoupler VI and a receiving optocoupler V2. When pulse signals are applied to input X, the LED of optocoupler VI periodically emits light; as a result, the output transistor of this optocoupler periodically saturates and the resistance between points a and b drops from several hundred kilo-ohms to several tens of ohms.

When the output transistor of the transmitting optocoupler is turned on, the current from the positive pole of the source U2 passes through the LED of the optocoupler V2, line (points a and b) and returns to the negative pole of this source. Source U2 is performed isolated from source U3.

If the output transistor of the transmitting optocoupler is turned off, then no current flows through the source circuit U2. The signal X" at the output of optocoupler V2 is close to zero if its LED is on, and close to +4 V if this LED is off. Thus, when X==0, the LEDs of the transmitting and receiving optocouplers are on and, therefore, X"==0. When X==1 both LEDs are off and X"==1.

Optocoupler isolation can significantly increase the noise immunity of the communication channel and ensure the transmission of information over distances of the order of hundreds of meters. Diodes connected to the transmitting and receiving optocouplers serve to protect them from reverse voltage surges. The resistor circuit connected to the source U2 serves to set the current in the line and limit the current through the LED of the receiving optocoupler.

The current in the line according to the IRPS interface can be selected equal to 20 or 40 mA. When choosing resistor values, you need to take into account the ohmic resistance of the communication line. Circuit with an “active transmitter” (Fig. 4.17, b) differs from the previous one in that the power supply for the U2 line is located on the transmitter side. This does not provide any advantages - both circuits are essentially the same and are so-called "current loops".

The recommendations given in this chapter may seem too harsh to a novice circuit designer. The fight against interference seems to him like a “fight with a windmill,” and the lack of experience in designing devices of increased complexity creates the illusion that it is possible to create a working device without following any of the recommendations given.

Indeed, sometimes this is possible. There are even cases of serial production of such devices. However, in informal reviews of their work one can hear many interesting non-technical expressions, such as visit effect and some others, simpler and more understandable.

Switching power supplies, thyristor regulators, switches, powerful radio transmitters, electric motors, substations, any electrical discharges near power lines (lightning, welding machines, etc.) generate narrowband and broadband interference of various natures and spectral composition. This complicates the functioning of low-current sensitive equipment, introduces distortions into measurement results, causes failures and even failure of both instrument components and entire equipment complexes.

In symmetrical electrical circuits (ungrounded circuits and circuits with a grounded midpoint), antiphase interference manifests itself in the form of symmetrical voltages (at the load) and is called symmetrical; in foreign literature it is called “differential mode interference”. Common mode interference in a symmetrical circuit is called asymmetrical or common mode interference.

Symmetrical line interference usually predominates at frequencies up to several hundred kHz. At frequencies above 1 MHz, asymmetric interference predominates.

A fairly simple case is narrowband interference, the elimination of which comes down to filtering the fundamental (carrier) frequency of the interference and its harmonics. A much more complex case is high-frequency impulse noise, the spectrum of which occupies a range of up to tens of MHz. Dealing with such interference is a rather difficult task.

Only a systematic approach will help eliminate strong complex interference, including a list of measures to suppress unwanted components of the supply voltage and signal circuits: shielding, grounding, correct installation of power and signal lines and, of course, filtering. A huge number of filter devices of various designs, quality factors, applications, etc. are produced and used all over the world.

Depending on the type of interference and area of ​​application, filter designs also differ. But, as a rule, the device is a combination of LC circuits forming filter cascades and P-type filters.

An important characteristic of a surge protector is the maximum leakage current. In power applications, this current can reach levels that are dangerous to humans. Based on leakage current values, filters are classified according to safety levels: applications that allow human contact with the device housing and applications where contact with the housing is undesirable. It is important to remember that the filter housing requires mandatory grounding.

TE-Connectivity builds on Corcom's more than 50 years of experience in the design and development of electromagnetic and RF filters to offer the widest range of devices for use in a variety of industries and applications. A number of popular series from this manufacturer are presented on the Russian market.

B series general purpose filters

Series B filters (Figure 1) are reliable and compact filters at an affordable price. A wide range of operating currents, good quality factor and a wide selection of connection types provide a wide range of applications for these devices.

Rice. 1.

Series B includes two modifications - VB and EB, the technical characteristics of which are given in Table 1.

Table 1. Main technical characteristics of B series network filters

The electrical circuit of the filter is shown in Figure 2.

Rice. 2.

The attenuation of the interference signal in dB is shown in Figure 3.

Rice. 3.

T series filters

Filters in this series (Figure 4) are high-performance radio frequency filters for power circuits of switching power supplies. The advantages of the series are excellent suppression of anti-phase and common-mode interference, compact dimensions. Low leakage currents allow the T series to be used in low power consumption applications.

Rice. 4.

The series includes two modifications - ET and VT, the technical characteristics of which are given in Table 2.

Table 2. Main technical characteristics of T series network filters

The electrical circuit of the T series filter is shown in Figure 5.

Rice. 5.

The attenuation of the interference signal in dB when the line is loaded onto a 50 Ohm matching resistor is shown in Figure 6.

Rice. 6.

K series filters

K series filters (Figure 7) are general purpose radio frequency power filters. They are intended for use in power circuits with high-resistance loads. Excellent for cases where the line is subject to pulsed, continuous and/or pulsating radio frequency interference. Models with the EK index meet the requirements of standards for use in portable devices and medical equipment.

Rice. 7.

Filters with index C are equipped with a choke between the housing and the ground wire. The main electrical parameters of the K series network filters are given in Table 3.

Table 3. Main electrical parameters of K series network filters

The electrical circuit of the K series filter is shown in Figure 8.

Rice. 8.

The attenuation of the interference signal in dB when the line is loaded onto a 50 Ohm matching resistor is shown in Figure 9.

Rice. 9.

EMC series filters

Filters in this series (Figure 10) are compact and efficient two-stage RF power filters. They have a number of advantages: a high coefficient of attenuation of common-mode interference in the low-frequency region, a high coefficient of attenuation of anti-phase interference, and compact dimensions. The EMC series is focused on use in devices with switching power supplies.

Rice. 10.

The main technical characteristics are given in Table 4.

Table 4. Basic electrical parameters of EMC series network filters

The electrical circuit of the EMC series filter is shown in Figure 11.

Rice. 11.

The attenuation of the interference signal in dB when the line is loaded onto a 50 Ohm matching resistor is shown in Figure 12.

Rice. 12.

EDP ​​series filters

2. Corcom Product Guide, General purpose RFI filters for high impedance loads at low current B Series, TE Connectivity, 1654001, 06/2011, p. 15

3. Corcom Product Guide, PC board mountable general purpose RFI filters EBP, EDP & EOP series, TE Connectivity, 1654001, 06/2011, p. 21

4. Corcom Product Guide, Compact and cost-effective dual stage RFI power line filters EMC Series, TE Connectivity, 1654001, 06/2011, p. 24

5. Corcom Product Guide, Single phase power line filter for frequency converters FC Series, 1654001, 06/2011, p. 30

6. Corcom Product Guide, General purpose RFI power line filters - ideal for high-impedance loads K Series, 1654001, 06/2011, p. 49

7. Corcom Product Guide, High performance RFI power line filters for switching power supplies T Series, 1654001, 06/2011, p. 80

8. Corcom Product Guide, Compact low-current 3-phase WYE RFI filters AYO Series, 1654001, 06/2011, p. 111.

Obtaining technical information, ordering samples, delivery - e-mail:

Network and signal EMI/RFI filters from TE Connectivity. From board to industrial installation

Company TE Connectivity occupies a leading position in the world in the development and production of surge protectors for effective suppression of electromagnetic and radio frequency interference in electronics and industry. The model range includes more than 70 series of devices for filtering both power circuits from external and internal sources, and signal circuits in a wide range of applications.

The filters have the following design options: miniature for installation on a printed circuit board; cabinets of various sizes and types of connection of supply lines and load lines; in the form of ready-made power connectors and communication connectors for network and telephone equipment; industrial, made in the form of ready-made industrial cabinets.

Surge filters are produced for AC and DC applications, single- and three-phase networks, covering the range of operating currents 1...1200 A and voltages 120/250/480 VAC, 48...130 VDC. All devices are characterized by a low voltage drop - no more than 1% of the operating voltage. The leakage current, depending on the power and design of the filter, is 0.2...8.0 mA. The average frequency range for the series is 10 kHz...30 MHz. Series AQ designed for a wider frequency range: 10 kHz...1 GHz. Expanding the applications of its products, TE Connectivity produces filters for low and high impedance load circuits. For example, high-impedance filters of the series EP, H, Q, R And V for low impedance loads and low impedance series B, EC, ED, EF, G, K, N, Q, S, SK, T, W, X, Y And Z for high impedance loads.

Communication connectors with built-in signal filters are available in shielded, paired and low-profile designs.

Each filter produced by TE Connectivity undergoes double testing: at the assembly stage and already in the form of a finished product. All products comply with international quality and safety standards.

pulse switching control. A typical block diagram of PIP inclusion in the reception path of a KB transceiver is shown in Fig. 1.

The pulse interference signal received at the input of the noise suppressor is amplified at node A2 and then detected by pulse detector U2. Adjusting the detector response threshold allows you to optimize the suppressor's performance. Pointed pulses from the output of node U2 turn on the rectangular pulse shaper G1, which controls the operation of the key stage S1, located in the signal path of the receiving device. In Fig. Figure 2 shows one of the first published PIP diagrams.

The impulse noise suppressor itself is made on transistors VT2-VT4 and diodes VD1-VD3. The stage on VT2 is an IF amplifier. A pulse detector is assembled on diode VD1. The cascade on transistor VT3, together with diodes VD2, VD3, forms rectangular pulses that control the electronic switch on transistor VT4.

The passage in the signal path in this case is interrupted due to the fact that the output of the cascade on transistor VT1 (IF amplifier) ​​during PIP operation is short-circuited (at high frequency) to the common wire.

Despite its simplicity, the assembly assembled according to the diagram in Fig. 2, works well. By changing the data of the oscillating circuit, this PIP can be used in receivers with an intermediate frequency from 0.5 to 9 MHz.

The transistors shown in the diagram can be replaced with any of the KP306 (VT1, VT2) and KPZ0Z (VT3, VT4) series. Instead of 1N9I4 diodes, you can use any of the KD522 series, instead of 1N34A from the D311 series.

The cascade in which the signal is interrupted is an important element of the PIP and largely determines the quality of its operation. The attenuation of the signal when passing through this stage should not exceed 3 dB and at the same time, when the signal path is opened, reach 80 dB or more. In addition, the switching control pulses that arrive at this stage have an amplitude of several volts and should not penetrate the signal path, where the level of the useful signal can be measured in microvolts. To this it is necessary to add the following: since the PIP is installed before the main selection filter, it must withstand high-level signals and not cause nonlinear effects.

This problem was successfully solved by G3PDM

l]. The key stage he developed for the noise suppressor (Fig. 3) is made on a field-effect transistor VT1. The resistance between its source and drain, depending on the control voltage applied to the gate, varies from 100 ohms to several megaohms. Switching pulses here can penetrate into the signal path through the gate-source capacitance (its value is 5...30 pF). To neutralize its effect, a control pulse in antiphase is supplied to the output circuit of the cascade through the capacitor SZ, by adjusting which it is possible to almost completely eliminate switching noise. When manufacturing a cascade, transistor 2N3823 can be replaced with KPZ0ZA, 2N4289 with KT361A.

Dissatisfaction with the quality of work of the key cascade in traditional PIP was the reason for further searches. W5QJR proposed that in KB receivers with double frequency conversion the control pulse should be applied not to the key stage, but to the second local oscillator. If sufficiently narrowband filters are installed in the first and second IF paths, then shifting the frequency of the second local oscillator by several kilohertz will lead to the fact that the signal and noise will no longer fall into the passband of the second filter, i.e., the signal path will be open. Since it is often reduced by only a few kilohertz, the normal operation of the local oscillator is maintained, there are no non-stationary transient processes, and with them switching interference.

The quality of work of this PIP is characterized by the following example. When installing a KB radio receiver in a car, reception without PIP was impossible, since powerful pulse interference from the ignition system completely clogged the signals of amateur stations. When the PIP was turned on, interference from the ignition system practically did not interfere with reception. In the W5QJR interference suppressor, a separate 38.8 MHz pulsed superheterodyne receiver is connected to the main receiver antenna. An amplified pulse signal at a frequency of 10.7 MHz is detected and supplied to the delay unit for controlling the commutation of the pulse and adjusting its duration. Part of the circuit diagram of this PIP is shown in Fig. 4.

A pulse detector is made on diode VD1. Cascades on VTI-VT3 transistors are included in the control signal generation unit. Logic elements DD1.1-DD1.4 form rectangular pulses supplied to a varicap connected to the local oscillator circuit, the frequency of which is shifted to the side.

Resistor R13 regulates the delay time of control pulses, and resistor R14 regulates their duration. Transistors VTI-VT3 can be any of the KT316 series, diode VD1 - any of the KD522 series, VD2 - D814A; DD1 - K561LE5.

Due to the fact that installation of the PIP developed by W5QJR is possible only in HF receivers that have fixed first and second IFs, it is natural that the search for the most acceptable option for a pulse noise suppressor continued. This was greatly facilitated by the appearance on the amateur HF bands of strong periodic interference, reminiscent of the knocking of a woodpecker. Since the strength of this interference is often up to S9+20 dB, it causes a lot of trouble for shortwave operators around the world.

Observations of the “woodpecker” and measurements of its parameters, given by VK1DN, showed that, unlike conventional pulsed interference (they have a pulse duration of 0.5...1 μs), this interference is longer (15 ms), repetition period is 10, sometimes 16 and much less frequently 20 and 32 Hz, its front and fall are not so steep, and the amplitude of the pulses arriving at a given moment can differ significantly from the previous ones.

This leads to the fact that not all impulse noise arriving at the receiver input triggers the PIP

and they freely penetrate the reception path. Knowing the quantitative characteristics of the “woodpecker” pulse, it is easy to conclude: in order to improve the performance of the interference suppressor, it is necessary to increase the gain in the pulsed interference receiving path, and also lengthen the switching pulse to 15 ms.

In Fig. Figure 5 shows a PIP, the development of which took into account the above considerations. The useful signal from the output of the mixer is fed to an IF amplifier assembled on field-effect transistors VT2 and VT3, and then through a switching stage on pulsed diodes VD1-VD4 is supplied to a quartz filter.

From the output of the mixer, through the source follower on transistor VT1, the IF signal branches into the pulse noise amplification path, which uses the DA1 microcircuit, which is part of a superheterodyne AM receiver (before the detector).

Its converter lowers the frequency of the incoming signal from 9 to 2 MHz. The detected interference pulse arrives through the source follower on transistor VT5 to the trigger unit assembled on transistor VT6.

The variable resistor R14 is used to regulate the PIP response threshold during operation, depending on the broadcast situation. Microcircuit DD1 generates a control pulse, which is supplied to the key stage through an inverting amplifier on transistor VT4. The PIP described by DJ2LR can be installed in a receiver having an IF from 3 to 40 MHz. In this case, you only need to use the corresponding circuits at the input of the DA1 chip. Only the design of the key cascade is critical in manufacturing. It requires careful shielding and symmetrical arrangement of parts for better balancing and decoupling. When repeating a node, transistors of the KPZOZ series, VT2, VT3 - KP903 series, VT4 - KT316 series, VT6 - KT361 series can be used as elements VT1, VT5. DA1 - K174ХА2, DD1 - K155AGZ.

The measurement data provided indicates the high parameters of the created unit. The signal attenuation at the moment the signal path is opened exceeds 80 dB. The value characterizing the upper limit of the dynamic range is +26 dBm. And most importantly, it was possible to completely get rid of the impulse noise created by the “woodpecker,” which made it possible to receive even very weak signals from DX stations. The article concludes that installing this PIP in high-end receiving devices will not lead to a deterioration in their dynamic range.

Measurements of the parameters of impulse noise from the “woodpecker”, which were reported by VK1DN, showed that these oscillations are very stable - with an accuracy of 10~5. This allows you to start the control pulse generation unit not with incoming interference, but with a signal from a local generator. Naturally, it must be highly stable and be able to adjust the output signal taking into account the phase of incoming signals.

In Fig. Figure 6 shows part of the PIP diagram developed by VK1DN. Trimmer resistors R3 and R6 adjust the control pulse, achieving the best suppression of interference.

Since the formation of the trigger pulse no longer actually depends on the construction of the KB receiver, VK1DN considers it possible to include a cascade switch in the LF path of the receiver. Despite the fact that it is not possible to completely get rid of interference and, in addition, the AGC system “breathes,” there is still a positive effect. The node can use the K555TL2 microcircuit, the KT316 series transistor, and the KD522 series diodes.

In Fig. Figure 7 shows the key stage of the low-frequency PIP and its triggering unit. Since the VK1DN uses a field-effect transistor as a switch, it is natural that it encountered the problem of control pulses “creeping” into the signal path, as mentioned at the beginning of the article. He solved it in his own way. It turned out that this interference can be significantly reduced by reducing the steepness of the front and fall of the control pulses.

To do this, at the output of the buffer stage on the operational amplifier DA1, separating the generator of these pulses from the rest of the device, a high-capacity capacitor C1 was installed - 33 μF. It, together with elements C2 and VD1, forms a triangular pulse with an amplitude of 9 V from a rectangular pulse. Transistor VT1 turns off when the voltage at its base is 7 V (for transistor MPF102). The node can use a K140UD7 microcircuit, a KPZ0Z series transistor, and a KD522 series diode.

According to VK1DN, it is advisable to power digital stages from a separate source in order to avoid interference from penetrating into the low-frequency path. The control signal to the low-frequency PIP should be supplied from the output of element DD1.5, and to the high-frequency one from transistor VT1 (see Fig. 6). This must be done so that the control pulse has the desired polarity.

Since the original source does not contain information about how the key stage in the VK1DN HF PIP was implemented, attention should be paid to this when repeating or experimenting.

S. Kazakov

Literature:

2. Van Zant F. Solid state noise blanker. - QST, 1971, No. 7, p. 20,

3. Hawker P. Technical topics.- Radio communication, 1978, No. 12, p. 1025.

4. Nicholls D. Blankihg the woob-pecker.- Harn Radio, 1982, No. 1, p. 20.

5. Ronde U. Increasing Receiver Dynamie Range. - QST, 1980, No. 5, p. 16.

6. Nicholls D. Blanking the woobpecker. - Ham Radio, 1982, No. 3, p. 22.